4-Channel, 16-Bit, 200 kSPS
CONTROL LOGIC
&
CALIBRATION CIRCUITRY
PWRD
V1A
V1B
BIP
CAP
REF
V
DIGVANA
EXT/INT
DATACLK
R/C
CS
SYNC
DGND
BUSY
WR2
WR1
A1A0
AGND2AGND1
REF
BUFF
2.5V
REFERENCE
AD974
4 TO 1
MUX
+
LATCH
EN
V2A
V2B
V3A
V3B
V4A
V4B
SWITCHED
CAP ADC
SERIAL
INTERFACE
16
DATA
RESISTIVE
NETWORK
RESISTIVE
NETWORK
RESISTIVE
NETWORK
RESISTIVE
NETWORK
CLOCK
a
FEATURES
Fast 16-Bit ADC with 200 kSPS Throughput
Four Single-Ended Analog Input Channels
Single +5 V Supply Operation
Input Ranges: 0 V to +4 V, 0 V to +5 V and ⴞ10 V
120 mW Max Power Dissipation
Power-Down Mode 50 W
Choice of External or Internal 2.5 V Reference
On-Chip Clock
Power-Down Mode
GENERAL DESCRIPTION
The AD974 is a four-channel, data acquisition system with a
serial interface. The part contains an input multiplexer, a highspeed 16-bit sampling ADC and a +2.5 V reference. All of this
operates from a single +5 V power supply that also has a powerdown mode. The part will accommodate 0 V to +4 V, 0 V to
+5 V or ±10 V analog input ranges.
The interface is designed for an efficient transfer of data while
requiring a low number of interconnects.
The AD974 is comprehensively tested for ac parameters such as
SNR and THD, as well as the more traditional parameters of
offset, gain and linearity.
The AD974 is fabricated on Analog Devices’ BiCMOS process,
which has high performance bipolar devices along with CMOS
transistors.
The AD974 is available in 28-lead DIP, SOIC and SSOP
packages.
Data Acquisition System
AD974
FUNCTIONAL BLOCK DIAGRAM
PRODUCT HIGHLIGHTS
1. The AD974 is a complete data acquisition system combining
a four-channel multiplexer, a 16-bit sampling ADC and a
+2.5 V reference on a single chip.
2. The part operates from a single +5 V supply and also has a
power-down feature.
3. Interfacing to the AD974 is simple with a low number of
interconnect signals.
4. The AD974 is comprehensively specified for ac parameters
such as SNR and THD, as well as dc parameters such as
linearity and offset and gain errors.
REV. A
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 781/329-4700 World Wide Web Site: http://www.analog.com
Fax: 781/326-8703 © Analog Devices, Inc., 1999
AD974–SPECIFICATIONS
(–40ⴗC to +85ⴗC, fS = 200 kHz, V
DIG
= V
= +5 V, unless otherwise noted)
ANA
A Grade B Grade
Parameter Conditions Min Typ Max Min Typ Max Units
RESOLUTION 16 16 Bits
ANALOG INPUT
Voltage Range ±10 V, 0 V to +4 V, 0 V to +5 V (See Table I)
Impedance Channel On or Off (See Table I)
Sampling Capacitance 40 40 pF
THROUGHPUT SPEED
Complete Cycle
(Acquire and Convert) 5 5 µs
Throughput Rate 200 200 kHz
DC ACCURACY
Integral Linearity Error ±3 ±2.0 LSB
Differential Linearity Error –2 +3 –1 +1.75 LSB
No Missing Codes 15 16 Bits
Transition Noise
Full-Scale Error
2
3
Internal Reference ±0.5 ±0.25 %
1.0 1.0 LSB
Full-Scale Error Drift Internal Reference ±7 ±7 ppm/°C
Full-Scale Error Ext. REF = +2.5 V ±0.5 ±0.25 %
Full-Scale Error Drift Ext. REF = +2.5 V ±2 ±2 ppm/°C
Bipolar Zero Error Bipolar Range ±10 ±10 mV
Bipolar Zero Error Drift Bipolar Range ±2 ±2 ppm/°C
Unipolar Zero Error Unipolar Ranges ±10 ±10 mV
Unipolar Zero Error Drift Unipolar Ranges ±2 ±2 ppm/°C
Channel-to-Channel Matching ±0.1 ±0.05 % FSR
Recovery to Rated Accuracy
After Power-Down
4
2.2 µF to CAP 1 1 ms
Power Supply Sensitivity
V
= V
ANA
DIG
= V
D
AC ACCURACY
Spurious Free Dynamic Range f
V
= 5 V ± 5% ±8 ±8 LSB
D
= 20 kHz 90 96 dB
IN
5
Total Harmonic Distortion fIN = 20 kHz –90 –96 dB
Signal-to-(Noise+Distortion) f
= 20 kHz 83 85 dB
IN
–60 dB Input 27 28 dB
Signal-to-Noise f
Channel-to-Channel Isolation f
Full Power Bandwidth
6
= 20 kHz 83 85 dB
IN
= 20 kHz –110 –100 –110 –100 dB
IN
1 1 MHz
–3 dB Input Bandwidth 2.7 2.7 MHz
SAMPLING DYNAMICS
Aperture Delay 40 40 ns
Transient Response Full-Scale Step 1 1 µs
Overvoltage Recovery
7
150 150 ns
REFERENCE
Internal Reference Voltage 2.48 2.5 2.52 2.48 2.5 2.52 V
Internal Reference Source Current 1 1 µA
External Reference Voltage Range
for Specified Linearity 2.3 2.5 2.7 2.3 2.5 2.7 V
External Reference Current Drain Ext. REF = +2.5 V 100 100 µA
DIGITAL INPUTS
Logic Levels
V
IL
V
IH
I
IL
I
IH
–0.3 +0.8 –0.3 +0.8 V
+2.0 V
+ 0.3 +2.0 V
DIG
+ 0.3 V
DIG
±10 ±10 µA
±10 ±10 µA
1
REV. A–2–
AD974
A Grade B Grade
Parameter Conditions Min Typ Max Min Typ Max Units
DIGITAL OUTPUTS
Data Format Serial 16 Bits
Data Coding Straight Binary
I
V
OL
V
OH
Output Capacitance High-Z State 15 15 pF
Leakage Current High-Z State
POWER SUPPLIES
Specified Performance
V
DIG
V
ANA
I
DIG
I
ANA
Power Dissipation
PWRD LOW 120 120 mW
PWRD HIGH 50 50 µW
TEMPERATURE RANGE
Specified Performance T
NOTES
1
LSB means Least Significant Bit. With a ±10 V input, one LSB is 305 µV.
2
Typical rms noise at worst case transitions and temperatures.
3
Full-Scale Error is expressed as the % difference between the actual full-scale code transition voltage and the ideal full-scale transition voltage, and includes the effect
of offset error. For bipolar input, the Full-Scale Error is the worst case of either the –Full-Scale or +Full-Scale code transition voltage errors. For unipolar input
ranges, Full-Scale Error is with respect to the +Full-Scale code transition voltage.
4
External 2.5 V reference connected to REF.
5
All specifications in dB are referred to a full-scale ±10 V input.
6
Full-Power Bandwidth is defined as full-scale input frequency at which Signal-to-(Noise + Distortion) degrades to 60 dB, or 10 bits of accuracy.
7
Recovers to specified performance after a 2 × FS input overvoltage.
Specifications subject to change without notice.
= 1.6 mA +0.4 +0.4 V
SINK
I
= 500 µA+4 +4 V
SOURCE
V
= 0 V to V
OUT
DIG
±5 ±5 µA
+4.75 +5 +5.25 +4.75 +5 +5.25 V
+4.75 +5 +5.25 +4.75 +5 +5.25 V
4.5 4.5 mA
14 14 mA
MIN
to T
MAX
–40 +85 –40 +85 °C
TIMING SPECIFICATIONS
(fS = 200 kHz, V
DIG
= V
= +5 V, –40ⴗC to +85ⴗC)
ANA
Parameter Symbol Min Typ Max Units
Convert Pulsewidth t
R/C, CS to BUSY Delay t
BUSY LOW Time t
BUSY Delay after End of Conversion t
Aperture Delay t
Conversion Time t
Acquisition Time t
Throughput Time t
R/C Low to DATACLK Delay t
DATACLK Period t
DATA Valid Setup Time t
DATA Valid Hold Time t
EXT. DATACLK Period t
EXT. DATACLK HIGH t
EXT. DATACLK LOW t
R/C, CS to EXT. DATACLK Setup Time t
R/C to CS Setup Time t
EXT. DATACLK to SYNC Delay t
EXT. DATACLK to DATA Valid Delay t
CS to EXT. DATACLK Rising Edge Delay t
Previous DATA Valid after CS, R/C Low t
BUSY to EXT. DATACLK Setup Time t
Final EXT. DATACLK to BUSY Rising Edge t
A0, A1 to WR1, WR2 Setup Time t
A0, A1 to WR1, WR2 Hold Time t
WR1, WR2 Pulsewidth t
Specifications subject to change without notic e.
1
2
3
4
5
6
7
+ t
6
8
9
10
11
12
13
14
15
16
17
18
19
20
21
22
23
24
25
7
50 ns
100 ns
4.0 µs
50 ns
40 ns
3.8 4.0 µs
1.0 µs
5 µs
220 ns
220 ns
50 ns
20 ns
66 ns
20 ns
30 ns
20 t12 + 5 ns
10 ns
15 66 ns
25 66 ns
10 ns
3.5 µs
5ns
1.7 µs
10 ns
10 ns
50 ns
REV. A –3–
AD974
WARNING!
ESD SENSITIVE DEVICE
TOP VIEW
(Not to Scale)
28
27
26
25
24
23
22
21
20
19
18
17
16
15
1
2
3
4
5
6
7
8
9
10
11
12
13
14
AD974
DGND
EXT/INT
PWRD
V
DIG
R/C
AGND2
REF
AGND1
V3A
V3B
V4A
CAP
BIP
V4B
SYNC
DATACLK
DATA
WR2
WR1
CS
BUSY
V2B
V2A
V1B
V1A
A1
A0
V
ANA
TO OUTPUT
PIN
C
L
100pF
I
OL
+1.4V
I
OH
1.6mA
500mA
ABSOLUTE MAXIMUM RATINGS
1
Analog Inputs
VxA, VxB . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±25 V
CAP . . . . . . . . . . . . . . . . +V
+ 0.3 V to AGND2 – 0.3 V
ANA
REF . . . . . . . . . . . . . . . . . . . . Indefinite Short to AGND2,
Momentary Short to V
ANA
Ground Voltage Differences
DGND, AGND1, AGND2 . . . . . . . . . . . . . . . . . . . ±0.3 V
Supply␣ Voltages
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +7 V
V
ANA
to V
V
DIG
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . +7 V
V
DIG
Digital Inputs . . . . . . . . . . . . . . . . . . . –0.3 V to V
Internal␣ Power␣ Dissipation
. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . ±7 V
ANA
2
DIG
+ 0.3 V
PDIP (N), SOIC (R), SSOP (RS) . . . . . . . . . . . . . 700 mW
Junction Temperature . . . . . . . . . . . . . . . . . . . . . . . . .+150°C
Storage Temperature Range N, R . . . . . . . . –65°C to +150°C
Lead Temperature Range
(Soldering␣ 10␣ sec) . . . . . . . . . . . . . . . . . . . . . . . . . .+300°C
NOTES
1
Stresses above those listed under Absolute Maximum Ratings may cause perma-
nent damage to the device. This is a stress rating only; functional operation of the
device at these or any other conditions above those indicated in the operational
section of this specification is not implied. Exposure to absolute maximum rating
conditions for extended periods may affect device reliability.
2
Specification is for device in free air:
28-Lead PDIP: θJA = 100°C/W, θJC = 31°C/W
28-Lead SOIC: θJA = 75°C/W, θJC = 24°C/W
28-Lead SSOP: θJA = 109°C/W, θJC = 39°C/W
PIN CONFIGURATION
SOIC, DIP AND SSOP
Figure 1. Load Circuit for Digital Interface Timing
ORDERING GUIDE
Temperature Package Package
Model Range Max INL Min S/(N+D) Description Options
AD974AN –40°C to +85°C ±3.0 LSB 83 dB 28-Lead Plastic DIP N-28B
AD974BN –40°C to +85°C ±2.0 LSB 85 dB 28-Lead Plastic DIP N-28B
AD974AR –40°C to +85°C ±3.0 LSB 83 dB 28-Lead SOIC R-28
AD974BR –40°C to +85°C ±2.0 LSB 85 dB 28-Lead SOIC R-28
AD974ARS –40°C to +85°C ±3.0 LSB 83 dB 28-Lead SSOP RS-28
AD974BRS –40°C to +85°C ±2.0 LSB 85 dB 28-Lead SSOP RS-28
CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the AD974 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.
–4–
REV. A
AD974
PIN FUNCTION DESCRIPTIONS
Pin No. Mnemonic Description
1 AGND1 Analog Ground. Used as the ground reference point for the REF pin.
2–5, 25–28 VxA, VxB Analog Input. Refer to Table I for input range configuration.
6 BIP Bipolar Offset. Connect VxA inputs to provide Bipolar input range.
7 CAP Reference Buffer Output. Connect a 2.2 µF tantalum capacitor between CAP and Analog
Ground.
8 REF Reference Input/Output. The internal +2.5 V reference is available at this pin. Alternatively an
external reference can be used to override the internal reference. In either case, connect a 2.2 µF
tantalum capacitor between REF and Analog Ground.
9 AGND2 Analog Ground.
10 R/C Read/Convert Input. Used to control the conversion and read modes. With CS LOW, a falling
edge on R/C holds the analog input signal internally and starts a conversion; a rising edge enables
the transmission of the conversion result.
11 V
12 PWRD Power-Down Input. When set to a logic HIGH, power consumption is reduced and conversions
13 EXT/INT Digital select input for choosing the internal or an external data clock. With EXT/INT tied LOW,
14 DGND Digital Ground.
15 SYNC Digital output frame synchronization for use with an external data clock (EXT/INT = Logic
16 DATACLK Serial data clock input or output, dependent upon the logic state of the EXT/INT pin. When
17 DATA The serial data output is synchronized to DATACLK. Conversion results are stored in an on-
18, 19 WR1, WR2 Multiplexer Write Inputs. These inputs are internally ORed to generate the mux latch inputs.
20 CS Chip Select Input. With R/C LOW, a falling edge on CS will initiate a conversion. With R/C
21 BUSY Busy Output. Goes LOW when a conversion is started, and remains LOW until the conversion is
22, 23 A1, A0 Address multiplexer inputs latched with the WR1, WR2 inputs.
DIG
Digital Power Supply. Nominally +5 V.
are inhibited. The conversion result from the previous conversion is stored in the onboard shift
register.
after initiating a conversion, 16 DATACLK pulses transmit the previous conversion result as
shown in Figure 3. With EXT/INT set to a Logic HIGH, output data is synchronized to an
external clock signal connected to the DATACLK input. Data is output as indicated in Figure 4
through Figure 9.
HIGH). When a read sequence is initiated, a pulse one DATACLK period wide is output
synchronous to the external data clock.
using the internal data clock (EXT/INT = Logic LOW), a conversion start sequence will initiate
transmission of 16 DATACLK periods. Output data is synchronous to this clock and is valid on
both its rising and falling edges (Figure 3). When using an external data clock (EXT/INT = Logic
HIGH), the CS and R/C signals control how conversion data is accessed.
chip register. The AD974 provides the conversion result, MSB first, from its internal shift regis-
ter. When using the internal data clock (EXT/INT = Logic LOW), DATA is valid on both the
rising and falling edges of DATACLK. Using an external data clock (EXT/INT = Logic HIGH)
allows previous conversion data to be accessed during a conversion (Figures 5, 7 and 9) or the
conversion result can be accessed after the completion of a conversion (Figures 4, 6 and 8).
The latch is transparent when WR1 and WR2 are tied low.
HIGH, a falling edge on CS will enable the serial data output sequence.
completed and the data is latched into the on-chip shift register.
24 V
REV. A
ANA
A1 A0 Data Available from Channel
00AIN 1
01AIN 2
10AIN 3
11AIN 4
Analog Power Supply. Nominally +5 V.
–5–
AD974
DEFINITION OF SPECIFICATIONS
INTEGRAL NONLINEARITY ERROR (INL)
Linearity error refers to the deviation of each individual code
from a line drawn from “negative full scale” through “positive
full scale.” The point used as “negative full scale” occurs 1/2 LSB
before the first code transition. “Positive full scale” is defined as
a level 1 1/2 LSB beyond the last code transition. The deviation
is measured from the middle of each particular code to the true
straight line.
DIFFERENTIAL NONLINEARITY ERROR (DNL)
In an ideal ADC, code transitions are 1 LSB apart. Differential
nonlinearity is the maximum deviation from this ideal value. It
is often specified in terms of resolution for which no missing
codes are guaranteed.
FULL-SCALE ERROR
The last + transition (from 011 . . . 10 to 011 . . . 11) should
occur for an analog voltage 1 1/2 LSB below the nominal full
scale (9.9995422 V for a ±10 V range). The full-scale error is
the deviation of the actual level of the last transition from the
ideal level.
BIPOLAR ZERO ERROR
Bipolar zero error is the difference between the ideal midscale
input voltage (0 V) and the actual voltage producing the midscale output code.
UNIPOLAR ZERO ERROR
In unipolar mode, the first transition should occur at a level
1/2 LSB above analog ground. Unipolar zero error is the deviation of the actual transition from that point.
SPURIOUS FREE DYNAMIC RANGE
The difference, in decibels (dB), between the rms amplitude of
the input signal and the peak spurious signal.
TOTAL HARMONIC DISTORTION (THD)
THD is the ratio of the rms sum of the first six harmonic components to the rms value of a full-scale input signal and is expressed in decibels.
SIGNAL TO (NOISE AND DISTORTION) (S/[N+D]) RATIO
S/(N+D) is the ratio of the rms value of the measured input
signal to the rms sum of all other spectral components below
the Nyquist frequency, including harmonics but excluding dc.
The value for S/(N+D) is expressed in decibels.
FULL POWER BANDWIDTH
The full power bandwidth is defined as the full-scale input frequency at which the S/(N+D) degrades to 60 dB, 10 bits of
accuracy.
APERTURE DELAY
Aperture delay is a measure of the acquisition performance, and
is measured from the falling edge of the R/C input to when the
input signal is held for a conversion.
TRANSIENT RESPONSE
The time required for the AD974 to achieve its rated accuracy
after a full-scale step function is applied to its input.
OVERVOLTAGE RECOVERY
The time required for the ADC to recover to full accuracy after
an analog input signal 150% of full-scale is reduced to 50% of
the full-scale value.
–6–
REV. A
AD974
CONVERSION CONTROL
The AD974 is controlled by two signals: R/C and CS. When
R/C is brought low, with CS low, for a minimum of 50 ns, the
input signal will be held on the internal capacitor array and a
conversion “n” will begin. Once the conversion process does
begin, the BUSY signal will go low until the conversion is complete. Internally, the signals R/C and CS are ORed together and
there is no requirement on which signal is taken low first when
initiating a conversion. The only requirement is that there be at
least 10 ns of delay between the two signals being taken low.
After the conversion is complete, the BUSY signal will return
high and the AD974 will again resume tracking the input signal.
Under certain conditions the CS pin can be tied Low and R/C
will be used to determine whether you are initiating a conversion or reading data. On the first conversion, after the AD974 is
powered up, the DATA output will be indeterminate.
Conversion results can be clocked serially, using either an
internal clock generated by the AD974 or an external clock.
The AD974 is configured for the internal data clock mode by
pulling the EXT/INT pin low. It is configured for the external
clock mode by pulling the EXT/INT pin high.
t
1
CS, R/C
A0, A1
INTERNAL DATA CLOCK MODE
The AD974 is configured to generate and provide the data clock
when the EXT/INT pin is held low. Typically CS will be tied
low and R/C will be used to initiate a conversion “n.” During
the conversion the AD974 will output 16 bits of data, MSB first,
from conversion “n-1” on the DATA pin. This data will be
synchronized with 16 clock pulses provided on the DATACLK
pin. The output data will be valid on both the rising and falling
edge of the data clock as shown in Figure 3. After the LSB has
been presented, the DATACLK pin will stay low until another
conversion is initiated.
In this mode, the digital input/output pins’ transitions are suitably positioned to minimize degradation on the conversion
result, mainly during the second half of the conversion process.
EXTERNAL DATA CLOCK MODE
The AD974 is configured to accept an externally supplied data
clock when the EXT/INT pin is held high. This mode of operation provides several methods by which conversion results can
be read. The output data from conversion “n-1” can be read
during conversion “n,” or the output data from conversion “n”
WR1, WR2
t25t
24
t
4
ACQUIRE
CONVERT
t
7
BUSY
MODE
t
t
2
t
5
ACQUIRE CONVERT
t
6
t
23
3
Figure 2. Basic Conversion Timing
t
8
R/C
DATACLK
DATA
BUSY
t
1
t
10
t
2
1
MSB
VALID
t
9
2 3 15 16
t
11
BIT 14
VALID
BIT 13
VALID
t
6
BIT 1
VALID
LSB
VALID
Figure 3. Serial Data Timing for Reading Previous Conversion Results with Internal Clock
(
CS
and EXT/
INT
Set to Logic Low)
REV. A
–7–
AD974
can be read after the conversion is complete. The external clock
can be either a continuous or discontinuous clock. A discontinuous clock can be either normally low or normally high when
inactive. In the case of the discontinuous clock, the AD974 can be
configured to either generate or not generate a SYNC output
(with a continuous clock a SYNC output will always be produced).
Each of the methods will be described in the following sections
and are illustrated in Figures 4 through 9. It should be noted
that all timing diagrams assume that the receiving device is
latching data on the rising edge of the external clock. If the
falling edge of DATACLK is used then, in the case of a discontinuous clock, one less clock pulse is required than shown in
Figures 4 through 7 to latch in a 16-bit word. Note that data is
valid on the falling edge of a clock pulse (for t
greater than t18)
13
and the rising edge of the next clock pulse.
The AD974 provides error correction circuitry that can correct
for an improper bit decision made during the first half of the
conversion cycle. Normally the occurrence of an incorrect bit
decision during a conversion cycle is irreversible. This error
occurs as a result of noise during the time of the decision or due
to insufficient settling time. As the AD974 is performing a
conversion it is important that transitions not occur on digital
input/output pins or degradation of the conversion result could
occur. This is particularly important during the second half of
the conversion process. For this reason it is recommended that
when an external clock is being provided it be a discontinuous
clock that is not toggling during the time that BUSY is low or,
more importantly, that it does not transition during the latter
half of BUSY low.
EXTERNAL DISCONTINUOUS CLOCK DATA READ
AFTER CONVERSION WITH NO SYNC OUTPUT
GENERATED
Figure 4 illustrates the method by which data from conversion
“n” can be read after the conversion is complete using a discontinuous external clock without the generation of a SYNC
output. After a conversion is complete, indicated by BUSY
returning high, the result of that conversion can be read while
CS is Low and R/C is high. In this mode CS can be tied low.
The MSB will be valid on the first falling edge and the second
rising edge of DATACLK. The LSB will be valid on the 16th
falling edge and the 17th rising edge of DATACLK. A minimum of 16 clock pulses are required for DATACLK if the
receiving device will be latching data on the falling edge of
DATACLK. A minimum of 17 clock pulses are required for
DATACLK if the receiving device will be latching data on the
rising edge of DATACLK.
The advantage of this method of reading data is that data is not
being clocked out during a conversion and therefore conversion
performance is not degraded.
When reading data after the conversion is complete, with the
highest frequency permitted for DATACLK (15.15 MHz), the
maximum possible throughput is approximately 195 kHz, and
not the rated 200 kHz.
t
12
t
13
t
EXT
DATACLK
R/C
BUSY
SYNC
DATA
0 123 141516
t
1
t
2
t
21
14
t
18
BIT 15
BIT 14 BIT 13 BIT 1
(MSB)
BIT 0
(LSB)
t
18
Figure 4. Conversion and Read Timing Using an External Discontinuous Data Clock
(EXT/
INT
Set to Logic High, CS Set to Logic Low)
–8–
REV. A
AD974
EXTERNAL DISCONTINUOUS CLOCK DATA READ
DURING CONVERSION WITH NO SYNC OUTPUT
GENERATED
Figure 5 illustrates the method by which data from conversion
“n-1” can be read during conversion “n” while using a discontinuous external clock, without the generation of a SYNC output. After a conversion is initiated, indicated by BUSY going
low, the result of the previous conversion can be read while CS
is low and R/C is high. In this mode CS can be tied low. The
MSB will be valid on the 1st falling edge and the 2nd rising edge of
DATACLK. The LSB will be valid on the 16th falling edge and
the 17th rising edge of DATACLK. A minimum of 16 clock
pulses are required for DATACLK if the receiving device will be
latching data on the falling edge of DATACLK. A minimum of
17 clock pulses are required for DATACLK if the receiving
device will be latching data on the rising edge of DATACLK.
In this mode the data should be clocked out during the first half
of BUSY so not to degrade conversion performance. This requires use of a 10 MHz DATACLK or greater, with data being
read out as soon as the conversion process begins.
EXTERNAL DISCONTINUOUS CLOCK DATA READ
AFTER CONVERSION WITH SYNC OUTPUT GENERATED
Figure 6 illustrates the method by which data from conversion “n” can be read after the conversion is complete using a
discontinuous external clock, with the generation of a SYNC
output. What permits the generation of a SYNC output is a
transition of DATACLK while either CS is high or while both
CS and R/C are low. After a conversion is complete, indicated
by BUSY returning high, the result of that conversion can be
read while CS is Low and R/C is high. In this mode CS can be
tied low. In Figure 6 clock pulse #0 is used to enable the generation of a SYNC pulse. The SYNC pulse is actually clocked
out approximately 40 ns after the rising edge of clock pulse #1.
The SYNC pulse will be valid on the falling edge of clock pulse
#1 and the rising edge of clock pulse #2. The MSB will be valid
on the falling edge of clock pulse #2 and the rising edge of clock
pulse #3. The LSB will be valid on the falling edge of clock
pulse #17 and the rising edge of clock pulse #18. The advantage of this method of reading data is that it is not being clocked
out during a conversion and therefore conversion performance is
not degraded.
When reading data after the conversion is complete, with the
highest frequency permitted for DATACLK (15.15 MHz), the
maximum possible throughput is approximately 195 kHz and
not the rated 200 kHz.
t
12
EXT
DATACLK
R/C
BUSY
SYNC
DATA
t
13
t
15
t
1
t
2
t
21
t
18
t
14
0
1 2 15 16
t
20
BIT 15
BIT 14
(MSB)
t
BIT 0
(LSB)
t
22
18
Figure 5. Conversion and Read Timing for Reading Previous Conversion Results During a Conversion
Using External Discontinuous Data Clock (EXT/
t
13
EXT
DATACLK
R/C
BUSY
SYNC
DATA
t
15
0
t
15
t
2
t
12
INT
Set to Logic High, CS Set to Logic Low)
t
12
t
14
123 18
t
15
t
17
t
18
BIT 15
(MSB)
417
t
18
BIT 14
BIT 0
(LSB)
REV. A
Figure 6. Conversion and Read Timing Using An External Discontinuous Data Clock
(EXT/
INT
Set to Logic High, CS Set to Logic Low)
–9–
AD974
EXTERNAL DISCONTINUOUS CLOCK DATA READ
DURING CONVERSION WITH SYNC OUTPUT
GENERATED
Figure 7 illustrates the method by which data from conversion
“n-1” can be read during conversion “n” while using a discontinuous external clock, with the generation of a SYNC output.
What permits the generation of a SYNC output is a transition of
DATACLK while either CS is High or while both CS and R/C
are low. In Figure 7 a conversion is initiated by taking R/C low
with CS tied low. While this condition exists a transition of
DATACLK, clock pulse #0, will enable the generation of a
SYNC pulse. Less then 83 ns after R/C is taken low the BUSY
output will go low to indicate that the conversion process has
t
12
13
12341718
t
15
t
1
t
17
t
14
t
12
t
18
BIT 15
(MSB)
EXT
DATACLK
R/C
BUSY
SYNC
DATA
t
0
t
15
t
2
begun. Figure 7 shows R/C then going high and after a delay of
greater than 15 ns (t
) clock pulse #1 can be taken high to
15
request the SYNC output. The SYNC output will appear approximately 40 ns after this rising edge and will be valid on the
falling edge of clock pulse #1 and the rising edge of clock pulse
#2. The MSB will be valid approximately 40 ns after the rising
edge of clock pulse #2 and can be latched off either the falling
edge of clock pulse #2 or the rising edge of clock pulse #3. The
LSB will be valid on the falling edge of clock pulse #17 and the
rising edge of clock pulse #18.
Data should be clocked out during the first half of BUSY to
avoid degrading conversion performance. This requires use of a
10 MHz DATACLK or greater, with data being read out as
soon as the conversion process begins.
t
22
t
20
t
18
BIT 14
BIT 0
(LSB)
Figure 7. Conversion and Read Timing for Reading Previous Conversion Results During a Conversion
Using External Discontinuous Data Clock (EXT/
INT
Set to Logic High, CS Set to Logic Low)
–10–
REV. A
AD974
EXTERNAL CONTINUOUS CLOCK DATA READ AFTER
CONVERSION WITH SYNC OUTPUT GENERATED
Figure 8 illustrates the method by which data from conversion
“n” can be read after the conversion is complete using a continuous external clock, with the generation of a SYNC output.
What permits the generation of a SYNC output is a transition of
DATACLK either while CS is high or while both CS and R/C are
low.
With a continuous clock the CS pin cannot be tied low as it
could be with a discontinuous clock. Use of a continuous clock,
while a conversion is occurring, can increase the DNL and
Transition Noise of the AD974.
After a conversion is complete, indicated by BUSY returning
high, the result of that conversion can be read while CS is low
t
12
EXT
DATACLK
CS
R/C
BUSY
SYNC
DATA
t
13
t
1
t
t
2
t
15
10
t
14
1 2 3 4 17 18
0
t
16
t
17
t
12
BIT 15
(MSB)
and R/C is high. In Figure 8 clock pulse #0 is used to enable the
generation of a SYNC pulse. The SYNC pulse is actually clocked
out approximately 40 ns after the rising edge of clock pulse #1.
The SYNC pulse will be valid on the falling edge of clock pulse
#1 and the rising edge of clock pulse #2. The MSB will be valid
on the falling edge of clock pulse #2 and the rising edge of clock
pulse #3. The LSB will be valid on the falling edge of clock
pulse #17 and the rising edge of clock pulse #18.
When reading data after the conversion is complete, with the
highest frequency permitted for DATACLK (15.15 MHz) the
maximum possible throughput is approximately 195 kHz and
not the rated 200 kHz.
t
19
t
18
BIT 14
BIT 0
(LSB)
t
18
Figure 8. Conversion and Read Timing Using an External Continuous Data Clock (EXT/
INT
Set to Logic High)
REV. A
–11–
AD974
EXTERNAL CONTINUOUS CLOCK DATA READ DURING
CONVERSION WITH SYNC OUTPUT GENERATED
Figure 9 illustrates the method by which data from conversion
“n-1” can be read during conversion “n” while using a continuous external clock with the generation of a SYNC output. What
permits the generation of a SYNC output is a transition of
DATACLK either while CS is high or while both CS and R/C
are low.
With a continuous clock the CS pin cannot be tied low as it
could be with a discontinuous clock. Use of a continuous clock
while a conversion is occurring can increase the DNL and
Transition Noise.
In Figure 9 a conversion is initiated by taking R/C low with CS
held low. While this condition exists a transition of DATACLK,
clock pulse #0, will enable the generation of a SYNC pulse. Less
then 83 ns after R/C is taken low the BUSY output will go low
t
12
t
13
t
EXT
DATACLK
CS
R/C
BUSY
SYNC
DATA
t
16
14
0123 18
t
15
t
1
t
2
t
17
t
12
BIT 15
(MSB)
to indicate that the conversion process has began. Figure 9
shows R/C then going high and after a delay of greater than
15 ns (t
), clock pulse #1 can be taken high to request the
15
SYNC output. The SYNC output will appear approximately
50 ns after this rising edge and will be valid on the falling edge
of clock pulse #1 and the rising edge of clock pulse #2. The
MSB will be valid approximately 40 ns after the rising edge of
clock pulse #2 and can be latched off either the falling edge of
clock pulse #2 or the rising edge of clock pulse #3. The LSB
will be valid on the falling edge of clock pulse #17 and the
rising edge of clock pulse #18.
Data should be clocked out during the 1st half of BUSY to
not degrade conversion performance. This requires use of a
10 MHz DATACLK or greater, with data being read out as
soon as the conversion process begins.
t
19
t
20
t
18
BIT 0
(LSB)
t
18
Figure 9. Conversion and Read Timing for Reading Previous Conversion Results During a Conversion
Using An External Continuous Data Clock (EXT/
INT
Set to Logic High)
–12–
REV. A
AD974
BIP AGND1 REF
CAP
VxA
VxB
AGND2
3kV
12kV
4kV
SWITCHED
CAP ADC
2.5V
REFERENCE
4kV
40pF
AD974
Table I. Analog Input Configuration
Input Voltage Connect Connect Input
Range VxA to VxB to Impedance
±10 V BIP V
0 V to +5 V V
0 V to +4 V V
IN
IN
IN
GND 6.0 kΩ
V
IN
Table II. Output Codes and Ideal Input Voltage
Description Analog Input Straight Binary
Full-Scale Range ±10 V 0 V to +5 V 0 V to +4 V
Least Significant Bit 305 µV 76 µV 61 µV
+Full Scale (FS – 1 LSB) +9.999695 V +4.999847 V +3.999939 V 1111 1111 1111 1111
Midscale 0 V +2.5 V +2 V 1000 0000 0000 0000
One LSB Below Midscale –305 µV +2.499924 V +1.999939 V 0111 1111 1111 1111
–Full Scale –10 V 0 V 0 V 0000 0000 0000 0000
13.7 kΩ
6.4 kΩ
Digital Input
ANALOG INPUTS
The AD974 is specified to operate with three full-scale analog
input ranges. Connections required for each of the eight analog
inputs, VxA and VxB and the resulting full-scale ranges, are
shown in Table I. The nominal input impedance for each analog input range is also shown. Table II shows the output codes
for the ideal input voltages of each of the analog input ranges.
The analog input section has a ±25␣ V overvoltage protection on
VxA and VxB. Since the AD974 has two analog grounds it is
important to ensure that the analog input is referenced to the
AGND1 pin, the low current ground. This will minimize any
problems associated with a resistive ground drop. It is also
important to ensure that the analog inputs are driven by a low
impedance source. With its primarily resistive analog input
circuitry, the ADC can be driven by a wide selection of general
purpose amplifiers.
To achieve the low distortion capability of the AD974 care
should be taken in the selection of the drive circuitry
op amp.
Figure 10 shows the simplified analog input section for the
AD974. Since the AD974 can operate with an internal or external reference, and three different analog input ranges, the fullscale analog input range is best represented with a voltage that
spans 0␣ V to V
across the 40 pF sampling capacitor. The on-
REF
chip resistors are laser trimmed to ratio match for adjustment of
offset and full-scale error using fixed external resistors.
Figure 10. Simplified Analog Input
REV. A
–13–
AD974
INPUT RANGE BASIC CONNECTIONS FOR AD974
BIP
VxA
V
610V
0V TO +5V
V
IN
2.2mF
2.2mF
IN
2.2mF
2.2mF
VxB
AGND1
CAP
+
AD974
REF
+
AGND2
BIP
VxA
VxB
AGND1
CAP
+
AD974
REF
+
AGND2
BIP
0V TO +4V
V
IN
2.2mF
2.2mF
VxA
VxB
AGND1
CAP
+
AD974
REF
+
AGND2
Figure 11. Analog Input Configurations
–14–
REV. A
AD974
BIP
VxA
VxB
AGND1
CAP
REF
AGND2
AD974
V
IN
C4
0.1mF
C2
2.2mF
+
V
ANA
TEMP V
OUT
GND
V
IN
AD780
6
4
2
3
C1
2.2mF
+
–
C3
1mF
+
–
+5V
0.1mF
–
OFFSET AND GAIN ADJUSTMENT
The AD974 is factory trimmed to minimize gain, offset and
linearity errors. There are no internal provisions to allow for any
further adjustment of offset error through external circuitry.
The reference of the AD974 can be adjusted as shown in Figure
12. This will allow the full-scale error of any one channel to be
adjusted to zero or will allow the average full-scale error of the
four channels to be minimized.
CAP
+
2.2mF
AD974
REF
AGND2
50kV
+5V
576kV
2.2mF
+
Figure 12. AD974 Full-Scale Trim
VOLTAGE REFERENCE
The AD974 has an on-chip temperature compensated bandgap
voltage reference that is factory trimmed to +2.5 V ± 20␣ mV.
The accuracy of the AD974 over the specified temperature
range is dominated by the drift performance of the voltage reference. The on-chip voltage reference is laser-trimmed to provide
a typical drift of 7␣ ppm/°C. This typical drift characteristic is
shown in Figure 13, which is a plot of the change in reference
voltage (in mV) versus the change in temperature—notice the
plot is normalized for zero error at +25°C. If improved drift perfor-
mance is required, an external reference such as the AD780
should be used to provide a drift as low as 3 ppm/°C. In order to
simplify the drive requirements of the voltage reference (internal
or external), an on-chip reference buffer is provided.
are taken to minimize any degradation in the ADC’s performance. Figure 14 shows the load regulation of the reference
buffer. Notice that this figure is also normalized so that there is
zero error with no dc load. In the linear region, the output imped-
ance at this point is typically 1 Ω. Because of this output imped-
ance, it is important to minimize any ac- or input-dependent
loads that will lead to increased distortion. Any dc load will
simply act as a gain error. Although the typical characteristic of
Figure 14 shows that the AD974 is capable of driving loads
greater than 15 mA, it is recommended that the steady state
current not exceed 2 mA.
dV ON CAP PIN – 10nV/DIV
SOURCE CAPABILITY SINK CAPABILITY
LOAD CURRENT – 5mA/DIV
Figure 14. CAP Pin Load Regulation
Using an External Reference
In addition to the on-chip reference, an external 2.5␣ V reference
can be applied. When choosing an external reference for a
16-bit application, however, careful attention should be paid to
noise and temperature drift. These critical specifications can
have a significant effect on the ADC performance.
Figure 15 shows the AD974 used in bipolar mode with the
AD780 voltage reference applied to the REF pin. The AD780
is a bandgap reference that exhibits ultralow drift, low initial
error and low output noise. For low power applications, the
AD780 provides a low quiescent current, high accuracy and low
temperature drift solution.
1mV/DIV
–55 25
DEGREES – Celsius
125
Figure 13. Reference Drift
The output of this buffer is provided at the CAP pin and is
available to the user; however, when externally loading the reference buffer, it is important to make sure that proper precautions
REV. A
Figure 15. External Reference to AD974 Configured for
±
10 V Input Range
–15–
AD974
AC PERFORMANCE
The AD974 is fully specified and tested for dynamic performance specifications. The ac parameters are required for signal
processing applications such as speech recognition and spectrum
analysis. These applications require information on the ADC’s
effect on the spectral content of the input signal. Hence, the
parameters for which the AD974 is specified include S/(N+D),
THD and Spurious Free Dynamic Range. These terms are
discussed in greater detail in the following sections.
As a general rule, it is recommended that the results from several conversions be averaged to reduce the effects of noise and
thus improve parameters such as S/(N+D) and THD. AC performance can be optimized by operating the ADC at its maximum sampling rate of 200 kHz and digitally filtering the resulting
bit stream to the desired signal bandwidth. By distributing noise
over a wider frequency range the noise density in the frequency
band of interest can be reduced. For example, if the required
input bandwidth is 50 kHz, the AD974 could be oversampled
by a factor of 4. This would yield a 6 dB improvement in the
effective SNR performance.
0
–10
–20
–30
–40
–50
–60
–70
–80
AMPLITUDE – dB
–90
–100
–110
–125
515 253545556575 8595
0 102030405060708090100
FREQUENCY – kHz
5280 POINT FFT
f
= 200kHz
SAMPLE
f
= 20kHz
IN
SNRD = 86.7dB
THD = 100.7dB
2.0
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
–2.0
0 5 10 15 20 25 30 35 40 45 50 55 60 66
100%
SAMPLES – K
Figure 17. INL Plot
2.0
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
–2.0
0 5 10 15 20 25 30 35 40 45 50 55 60 66
100%
SAMPLES – K
Figure 18. DNL Plot
Figure 16. FFT Plot
DC PERFORMANCE
The factory calibration scheme used for the AD974 compensates for bit weight errors that may exist in the capacitor array.
The mismatch in capacitor values is adjusted (using the calibration coefficients) during a conversion, resulting in excellent dc
linearity performance. Figures 17 and 18, respectively, show
typical INL and DNL plots for the AD974 at +25°C.
A histogram test is a statistical method for deriving an A/D
converter’s differential nonlinearity. A ramp input is sampled
by the ADC and a large number of conversions are taken at
each voltage level, averaged and then stored. The effect of
averaging is to reduce the transition noise by 1/n. If 64 samples
are averaged at each point, the effect of transition noise is
reduced by a factor of 8; i.e., a transition noise of 0.8 LSBs rms
is reduced to 0.1 LSBs rms. Theoretically the codes, during a
test of DNL, would all be the same size and therefore have an
equal number of occurrences. A code with an average number
of occurrences would have a DNL of “0.” A code that is
different from the average would have a DNL that was either
greater or less than zero LSB. A DNL of –1 LSB indicates that
there is a missing code present at the 16-bit level and that the
ADC exhibits 15-bit performance.
–16–
90
80
70
= 0dB
60
IN
50
40
SINAD (dB) FOR V
30
20
10
1 100010010
INPUT SIGNAL FREQUENCY – kHz
SNR+D (dB) FOR AD974
Figure 19. S/(N+D) vs. Input Frequency
REV. A
AD974
–80
–115
1 10 100 1000
–95
–100
–105
–110
–85
–90
10000
ACTIVE CHANNEL INPUT FREQUENCY – kHz
RESULTING AMPLITUDE ON SELECTED
CHANNEL (dB) WITH INPUT GROUNDED
ADJACENT CHANNELS,
WORST PAIR
NONADJACENT
CHANNELS
0
–130
12
–90
–100
–110
–120
–70
–80
FREQUENCY – kHz
dBFS
4 6 8 10 12 14 16 18 20
–60
–40
–50
–30
–10
–20
110
105
100
95
90
SFDR, S/N + D – dB
85
80
–75 150–50 –25
SFDR
THD
SNRD
0
25 50 75 100 125
TEMPERATURE – 8C
–80
–85
–90
–95
THD – dB
–100
–105
–110
Figure 20. AC Parameters vs. Temperature
DC CODE UNCERTAINTY
Ideally, a fixed dc input should result in the same output code
for repetitive conversions; however, as a consequence of unavoidable circuit noise within the wideband circuits of the ADC,
a range of output codes may occur for a given input voltage.
Thus, when a dc signal is applied to the AD974 input, and
10,000 conversions are recorded, the result will be a distribution
of codes as shown in Figure 21. This histogram shows a bell
shaped curve consistent with the Gaussian nature of thermal
noise. The histogram is approximately seven codes wide. The
standard deviation of this Gaussian distribution results in a code
transition noise of 1 LSB rms.
When used with an external reference, connected to the REF
pin and a 2.2 µF capacitor, connected to the CAP pin, the
power-up recovery time is typically 1 ms. This typical value of
1 ms for recovery time depends on how much charge has de-
cayed from the external 2.2 µF capacitor on the CAP pin and
assumes that it has decayed to zero. The 1 ms recovery time has
been specified such that settling to 16 bits has been achieved.
When used with the internal reference, the dominant time constant for power-up recovery is determined by the external capacitor on the REF pin and the internal 4K impedance seen at
that pin. An external 2.2 µF capacitor is recommended for the
REF pin.
CROSSTALK
The crosstalk between adjacent channels, nonadjacent channels
and worst-case adjacent channels is shown in Figures 22 to 24.
The worst-case crosstalk occurs between channels 1 and 2.
4000
3500
3000
2500
2000
1500
1000
500
0
–3 –2 –1
Figure 21. Histogram of 10,000 Conversions of a DC Input
POWER-DOWN FEATURE
The AD974 has analog and reference power-down capability
through the PWRD pin. When the PWRD pin is taken high,
the power consumption drops from a maximum value of
100 mW to a typical value of 50 µW. When in the power-
down mode the previous conversion results are still available in
the internal registers and can be read out providing it has not
already been shifted out.
REV. A
0
Figure 22. Crosstalk vs. Input Frequency (kHz)
1234
Figure 23. Adjacent Channel Crosstalk, Worst Pair
(8192 Point FFT; AIN 2 = 1.02 kHz, –0.1 dB; AIN 1 = AGND)
–17–
AD974
DR0
SCLK0
PF1
RFS0
ADSP-2181
DATA
DATACLK
R/C
EXT/INT
CS
AD974
PF0
OSCILLATOR
SPORT0 CNTRL REG = 03300F
0
–10
–20
–30
–40
–50
–60
dBFS
–70
–80
–90
–100
–110
–120
–130
12
4 6 8 10 12 14 16 18 20
FREQUENCY – kHz
Figure 24. Adjacent Channel Crosstalk, Worst Pair (8192
Point FFT; AIN 2 = 220 kHz, –0.1 dB; AIN 1 = AGND)
MICROPROCESSOR INTERFACING
The AD974 is ideally suited for traditional dc measurement
applications supporting a microprocessor, and ac signal processing applications interfacing to a digital signal processor. The
AD974 is designed to interface with a general purpose serial
port or I/O ports on a microcontroller. A variety of external
buffers can be used with the AD974 to prevent digital noise
from coupling into the ADC. The following sections illustrate
the use of the AD974 with an SPI equipped microcontroller and
the ADSP-2181 signal processor.
SPI INTERFACE
Figure 25 shows a general interface diagram between the
AD974 and an SPI equipped microcontroller. This interface
assumes that the convert pulses will originate from the microcontroller and that the AD974 will act as the slave device. The
convert pulse could be initiated in response to an internal timer
interrupt. The reading of output data, one byte at a time,
if necessary, could be initiated in response to the end-ofconversion signal (BUSY going high).
SDI
SCK
I/O PORT
IRQ
SPI
+5V
DATA
DATACLK
R/C
BUSY
EXT/INT
CS
AD974
Figure 25. AD974-to-SPI Interface
ADSP-2181 INTERFACE
Figure 26 shows an interface between the AD974 and the
ADSP-2181 Digital Signal Processor. The AD974 is configured
for the Internal Clock mode (EXT/INT = 0) and will therefore
act as the master device. The convert command is shown generated from an external oscillator in order to provide a low jitter
signal appropriate for both dc and ac measurements. Because
the SPORT, within the ADSP-2181, will be seeing a discontinuous external clock, some steps are required to ensure that the
serial port is properly synchronized to this clock during each
data read operation. The recommended procedure to ensure
this is as follows:
• Enable SPORT0 through the System Control register.
• Set the SCLK Divide register to zero.
• Setup PF0 and PF1 as outputs by setting bits 0 and 1 in
PFTYPE.
• Force RFS0 low through PF0. The Receive Frame Sync
signal has been programmed active high.
• Enable AD974 by forcing CS = 0 through PF1.
• Enable SPORT0 Receive Interrupt through the IMASK
register.
• Wait for at least one full conversion cycle of the AD974 and
throw away the received data.
• Disable the AD974 by forcing CS = 1 through PF1.
• Wait for a period of time equal to one conversion cycle.
• Force RFS0 high through PF0.
• Enable the AD974 by forcing CS = 0 through PF1.
The ADSP-2181 SPORT0 will now remain synchronized to the
external discontinuous clock for all subsequent conversions.
Figure 26. AD974-to-ADSP-2181 Interface
POWER SUPPLIES AND DECOUPLING
The AD974 has two power supply input pins. V
ANA
and V
provide the supply voltages to the analog and digital portions,
respectively. V
circuitry, and V
is the +5 V supply for the on-chip analog
ANA
is the +5 V supply for the on-chip digital
DIG
circuitry. The AD974 is designed to be independent of power
supply sequencing and thus free from supply voltage induced
latchup.
With high performance linear circuits, changes in the power
supplies can result in undesired circuit performance. Optimally,
well regulated power supplies should be chosen with less than
1% ripple. The ac output impedance of a power supply is a
complex function of frequency and will generally increase with
frequency. Thus, high frequency switching, such as that encountered with digital circuitry, requires the fast transient currents that most power supplies cannot adequately provide. Such
a situation results in large voltage spikes on the supplies. To
compensate for the finite ac output impedance of most supplies,
charge “reserves” should be stored in bypass capacitors. This
will effectively lower the supplies impedance presented to the
AD974 V
ANA
and V
pins and reduce the magnitude of these
DIG
spikes. Decoupling capacitors, typically 0.1␣ µF, should be placed
close to the power supply pins of the AD974 to minimize any
inductance between the capacitors and the V
–18–
ANA
and V
DIG
REV. A
DIG
pins.
AD974
The AD974 may be operated from a single +5␣ V supply.
When separate supplies are used, however, it is beneficial to
have larger (10␣ µF) capacitors placed between the logic supply
) and digital common (DGND), and between the analog
(V
DIG
supply (V
) and the analog common (AGND2). Addition-
ANA
ally, 10␣ µF capacitors should be located in the vicinity of the
ADC to further reduce low frequency ripple. In systems where
the device will be subjected to harsh environmental noise,
additional decoupling may be required.
GROUNDING
The AD974 has three ground pins; AGND1, AGND2 and
DGND. The analog ground pins are the “high quality” ground
reference points and should be connected to the system analog
common. AGND2 is the ground to which most internal ADC
analog signals are referenced. This ground is most susceptible to
current-induced voltage drops and thus must be connected with
the least resistance back to the power supply. AGND1 is the low
current analog supply ground and should be the analog common
for the external reference, input op amp drive circuitry and the
input resistor divider circuit. By applying the inputs referenced
to this ground, any ground variations will be offset and have a
minimal effect on the resulting analog input to the ADC. The
digital ground pin, DGND, is the reference point for all of the
digital signals that control the AD974.
The AD974 can be powered with two separate power supplies or
with a single analog supply. When the system digital supply is
noisy, or fast switching digital signals are present, it is recommended to connect the analog supply to both the V
ANA
and V
DIG
pins of the AD974 and the system supply to the remaining
digital circuitry. With this configuration, AGND1, AGND2 and
DGND should be connected back at the ADC. When there is
significant bus activity on the digital output pins, the digital and
analog supply pins on the ADC should be separated. This would
eliminate any high speed digital noise from coupling back to the
analog portion of the AD974. In this configuration, the digital
ground pin DGND should be connected to the system digital
ground and be separate from the AGND pins.
BOARD LAYOUT
Designing with high resolution data converters requires careful
attention to board layout and trace impedance is a significant
issue. A 1.22␣ mA current through a 0.5 Ω trace will develop a
voltage drop of 0.6 mV, which is 2 LSBs at the 16-bit level over
the 20␣ volt full-scale range. Ground circuit impedances should
be reduced as much as possible since any ground potential
differences between the signal source and the ADC appear as
an error voltage in series with the input signal. In addition to
ground drops, inductive and capacitive coupling needs to be
considered. This is especially true when high accuracy analog
input signals share the same board with digital signals. Thus, to
minimize input noise coupling, the input signal leads to V
IN
and
the signal return leads from AGND should be kept as short as
possible. In addition, power supplies should also be decoupled
to filter out ac noise.
Analog and digital signals should not share a common path.
Each signal should have an appropriate analog or digital return
routed close to it. Using this approach, signal loops enclose a
small area, minimizing the inductive coupling of noise. Wide
PC tracks, large gauge wire and ground planes are highly recommended to provide low impedance signal paths. Separate
analog and digital ground planes are also recommended with a
single interconnection point to minimize ground loops. Analog
signals should be routed as far as possible from high speed
digital signals and if absolutely necessary, should only cross
them at right angles.
In addition, it is recommended that multilayer PC boards be
used with separate power and ground planes. When designing
the separate sections, careful attention should be paid to the
layout.
REV. A
–19–
AD974
OUTLINE DIMENSIONS
Dimensions shown in inches and (mm).
28-Lead 300 Mil Plastic DIP
1.425 (38.195)
1.385 (35.179)
28
1
PIN 1
0.210
(5.33)
MAX
SEATING
PLANE
0.022 (0.558)
0.014 (0.356)
0.100 (2.54)
BSC
28-Lead Wide Body (SOIC)
0.7125 (18.10)
0.6969 (17.70)
28 15
(N-28B)
15
14
0.015 (0.381)
0.070 (1.77)
0.045 (1.15)
(R-28)
0.280 (7.11)
0.240 (6.10)
MIN
0.150 (3.81)
0.115 (2.92)
0.325 (8.25)
0.300 (7.62)
0.014 (0.356)
0.008 (0.204)
0.195 (4.95)
0.115 (2.93)
C3273a–0–5/99
0.0118 (0.30)
0.0040 (0.10)
0.311 (7.9)
0.301 (7.64)
0.078 (1.98)
0.068 (1.73)
0.008 (0.203)
0.002 (0.050)
0.2992 (7.60)
0.2914 (7.40)
PIN 1
0.0500
(1.27)
BSC
0.0192 (0.49)
0.0138 (0.35)
141
0.1043 (2.65)
0.0926 (2.35)
SEATING
PLANE
0.4193 (10.65)
0.3937 (10.00)
0.0125 (0.32)
0.0091 (0.23)
0.0291 (0.74)
0.0098 (0.25)
8°
0°
28-Lead Shrink Small Outline Package (SSOP)
(RS-28)
0.407 (10.34)
0.397 (10.08)
28 15
0.212 (5.38)
0.205 (5.21)
141
PIN 1
0.0256
(0.65)
BSC
0.015 (0.38)
0.010 (0.25)
0.07 (1.79)
0.066 (1.67)
SEATING
PLANE
0.009 (0.229)
0.005 (0.127)
8°
0°
x 45°
0.0500 (1.27)
0.0157 (0.40)
PRINTED IN U.S.A.
0.03 (0.762)
0.022 (0.558)
–20–
REV. A
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