Datasheet AD6653 Datasheet (ANALOG DEVICES)

A

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IF Diversity Receiver

FEATURES

SNR = 70.8 dBc (71.8 dBFS) in a 32.7 MHz BW at

70 MHz @ 150 MSPS

SFDR = 83 dBc to 70 MHz @ 150 MSPS

1.8 V analog supply operation

1.8 V to 3.3 V CMOS output supply or 1.8 V LVDS ou

tput supply Integer 1-to-8 input clock divider Integrated dual-channel ADC

Sample rates up to 150 MSPS

IF sampling frequencies to 450 MHz Internal ADC voltage reference Integrated ADC sample-and-hold inputs Flexible analog input range: 1 V p-p to 2 V p-p ADC clock duty cycle stabilizer 95 dB channel isolation/crosstalk

Integrated wideband digital downconverter (DDC)

32-bit, complex, numerically controlled oscillator (NCO) Decimating half-band filter and FIR filter Supports real and complex output modes Fast attack/threshold detect bits

Composite signal monitor Energy-saving power-down modes

FUNCTIONAL BLOCK DIAGRAM

AVDD FD[0:3]

AD6653

APPLICATIONS

Communications Diversity radio systems Multimode digital receivers (3G)

TD-SCDMA, WiMax, WCDMA,

CDMA20 I/Q demodulation systems Smart antenna systems General-purpose software radios Broadband data applications

PRODUCT HIGHLIGHTS

1. Integrated dual, 12-bit, 125 MSPS/150 MSPS ADC.

2. I

3. F

4. P

5. F

6. S

7. 3

00, GSM, EDGE, LTE

ntegrated wideband decimation filter and 32-bit

complex NCO.

ast overrange detect and signal monitor with serial output. roprietary differential input maintains excellent SNR

performance for input frequencies up to 450 MHz.

lexible output modes, including independent CMOS,

interleaved CMOS, IQ mode CMOS, and interleaved LVDS.

YNC input allows synchronization of multiple devices.

-bit SPI port for register programming and register readback.

DVDD DRVDD

FD BITS/T HRESHOLD

DETECT

VIN+A

VIN–A

VREF

SENSE

CML

RBIAS

VIN–B

VIN+B

NOTES

1. PIN NAMES ARE FOR THE CMO S PIN CONF IGURATIO N ONLY; SEE FIGURE 10 FOR LVDS PIN NAMES.

Rev. 0

Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Anal og Devices for its use, nor for any infringements of patents or ot her rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.

SHA

REF

SELECT

SHA ADC

MULTI-CHI P

AGND

SYNC

SYNC FD[0:3]B SMI

ADC

SIGNAL

MONITOR

FD BITS/THRESHOLD

DETECT

32-BIT

TUNING

NCO

SIGNAL MONITOR

DATA

I

DECIMATING

HB FILTER +

Q

DECIMATING

HB FILTER +

I

Figure 1.

AD6653

LP/HP

FIR

DIVIDE 1

f

/8

ADC

NCO

LP/HP

FIR

SIGNAL MO NITOR

INTERFACE

SMI SCLK/ PDWN

SMI

SDO/

OEB

SDFS

TO 8

DUTY

CYCLE

STABILIZER

PROGRAMMING DATA

SDIO/

DCS

SPI

SCLK/

DFS

GENERATION

CSB DRGND

CMOS/LV DS

DCO

CMOS

D11A

D0A

OUTPUT BUFFER

CLK+

CLK–

DCOA

DCOB

D11B

D0B

OUTPUT BUFFER

6708-001

AD6653

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REVISION HISTORY

11/07—Revision 0: Initial Version

Rev. 0 | Page 3 of 80

AD6653

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GENERAL DESCRIPTION

The AD6653 is a mixed-signal intermediate frequency (IF) receiver consisting of dual, 12-bit, 125 MSPS/150 MSPS ADCs and a wideband digital downconverter (DDC). The AD6653 is designed to support communications applications where low cost, small size, and versatility are desired.

The dual ADC core features a multistage, differential pipelined a

rchitecture with integrated output error correction logic. Each ADC features wide bandwidth differential sample-and-hold analog input amplifiers supporting a variety of user-selectable input ranges. An integrated voltage reference eases design considerations. A duty cycle stabilizer is provided to compensate for variations in the ADC clock duty cycle, allowing the converters to maintain excellent performance.

ADC data outputs are internally connected directly to the

tal downconverter (DDC) of the receiver, simplifying layout

digi and reducing interconnection parasitics. The digital receiver has two channels and provides processing flexibility. Each receive channel has four cascaded signal processing stages: a 32-bit frequency translator (numerically controlled oscillator (NCO)), a decimating half-band filter, a fixed FIR filter, and an f fixed-frequency NCO.

In addition to the receiver, DDC, the AD6653 has several functions tha

t simplify the automatic gain control (AGC) function in the system receiver. The fast detect feature allows fast overrange detection by outputting four bits of input level information with short latency.

ADC

/8

In addition, the programmable threshold detector allows

itoring of the incoming signal power using the four fast

mon detect bits of the ADC with low latency. If the input signal level exceeds the programmable threshold, the coarse upper threshold indicator goes high. Because this threshold indicator has low latency, the user can quickly turn down the system gain to avoid an overrange condition.

The second AGC-related function is the signal monitor. This block

lows the user to monitor the composite magnitude of the

al incoming signal, which aids in setting the gain to optimize the dynamic range of the overall system.

After digital processing, data can be routed directly to the two e

xternal 12-bit output ports. These outputs can be set from 1.8 V to 3.3 V CMOS or as 1.8 V LVDS. The CMOS data can also be output in an interleaved configuration at a double data rate, using only Port A.

The AD6653 receiver digitizes a wide spectrum of IF frequencies.

ach receiver is designed for simultaneous reception of the main

E channel and the diversity channel. This IF sampling architecture greatly reduces component cost and complexity compared with traditional analog techniques or less integrated digital methods. Flexible power-down options allow significant power savings, when desired.

Programming for setup and control is accomplished using a 3-bit

PI-compatible serial interface.

S

The AD6653 is available in a 64-lead LFCSP and is specified over t

he industrial temperature range of −40°C to +85°C.

Rev. 0 | Page 4 of 80

AD6653

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SPECIFICATIONS

ADC DC SPECIFICATIONS

AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, maximum sample rate, VIN = −1.0 dBFS differential input, 1.0 V internal reference, DCS enabled, unless otherwise noted.

Table 1.

AD6653BCPZ-125 AD6653BCPZ-150

Parameter Temperature

Min Typ Max Min Typ Max

RESOLUTION Full 12 12 Bits ACCURACY

No Missing Codes Full Guaranteed Guaranteed Offset Error Full ±0.3 ±0.6 ±0.2 ±0.6 % FSR Gain Error Full −3.9 −2.7 −0.7 −5.2 −3.2 −0.9 % FSR

MATCHING CHARACTERISTIC

Offset Error 25°C ±0.3 ±0.6 ±0.2 ±0.7 % FSR Gain Error 25°C ±0.1 ±0.7 ±0.2 ±0.7 % FSR

TEMPERATURE DRIFT

Offset Error Full ±19 ±17 ppm/°C Gain Error Full ±38 ±49 ppm/°C

INTERNAL VOLTAGE REFERENCE

Output Voltage Error (1 V Mode) Full ±5 ±18 ±5 ±18 mV Load Regulation @ 1.0 mA Full 7 7 mV

INPUT-REFERRED NOISE

VREF = 1.0 V 25°C 0.21 0.21 LSB rms

ANALOG INPUT

Input Span, VREF = 1.0 V Full 2 2 V p-p Input Capacitance

1

Full 8 8 pF VREF INPUT RESISTANCE Full 6 6 kΩ POWER SUPPLIES

Supply Voltage

AVDD, DVDD Full 1.7 1.8 1.9 1.7 1.8 1.9 V DRVDD (CMOS Mode) Full 1.7 3.3 3.6 1.7 3.3 3.6 V DRVDD (LVDS Mode) Full 1.7 1.8 1.9 1.7 1.8 1.9 V

Supply Current

2, 3

I

AVDD

2, 3

I

DVDD

2

I

(3.3 V CMOS) Full 20 24 mA

DRVDD

2

I

(1.8 V CMOS) Full 12 15 mA

DRVDD

2

I

(1.8 V LVDS) Full 57 57 mA

DRVDD

Full 390 440 mA

Full 270

689

320

785

POWER CONSUMPTION

DC Input Full 770 800 870 905 mW Sine Wave Input2 (DRVDD = 1.8 V) Full 1215 1395 mW Sine Wave Input2 (DRVDD = 3.3 V) Full 1275 1450 mW Standby Power

4

Full 77 77 mW

Power-Down Power Full 2.5 8 2.5 8 mW

1

Input capacitance refers to the effective capacitance between one differential input pin and AGND. See Figure 11 for the equivalent analog input structure.

2

Measured with a 9.7 MHz, full-scale sine wave input, NCO enabled with a frequency of 13 MHz, FIR filter enabled and the fS/8 output mix enabled with approximately

5 pF loading on each output bit.

3

The maximum limit applies to the combination of I

4

Standby power is measured with a dc input and with the CLK pin inactive (set to AVDD or AGND).

AVDD

and I

DVDD

currents.

Unit

mA

Rev. 0 | Page 5 of 80

AD6653

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ADC AC SPECIFICATIONS

AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, maximum sample rate, VIN = −1.0 dBFS differential input, 1.0 V internal reference, DCS enabled, NCO enabled, half-band filter enabled, FIR filter enabled, unless otherwise noted.

Table 2.

Parameter

SIGNAL-TO-NOISE-RATIO (SNR)

fIN = 2.4 MHz 25°C 71.0 70.9 dB fIN = 70 MHz 25°C 70.8 70.8 dB Full 69.8 69.4 dB fIN = 140 MHz 25°C 70.6 70.6 dB fIN = 220 MHz 25°C 70.2 70.0 dB

WORST SECOND OR THIRD HARMONIC

fIN = 2.4 MHz 25°C −85 −84 dBc fIN = 70 MHz 25°C −84 −83 dBc Full −74 −73 dBc fIN = 140 MHz 25°C −83 −82 dBc fIN = 220 MHz 25°C −81 −77 dBc

SPURIOUS-FREE DYNAMIC RANGE (SFDR)

fIN = 2.4 MHz 25°C 85 84 dBc fIN = 70 MHz 25°C 84 83 dBc Full 74 73 dBc fIN = 140 MHz 25°C 83 82 dBc fIN = 220 MHz 25°C 81 77 dBc

WORST OTHER HARMONIC OR SPUR

fIN = 2.4 MHz 25°C −92 −90 dBc fIN = 70 MHz 25°C −90 −87 dBc Full −82 −80 dBc fIN = 140 MHz 25°C −88 −83 dBc fIN = 220 MHz 25°C −84 −78 dBc

TWO-TONE SFDR

fIN = 29.12 MHz, 32.12 MHz (−7 dBFS) 25°C 85 85 dBc

fIN = 169.12 MHz, 172.12 MHz (−7 dBFS) 25°C 81 81 dBc CROSSTALK ANALOG INPUT BANDWIDTH 25°C 650 650 MHz

1

See Application Note AN-835, Understanding High Speed ADC Testing and Evaluation, for a complete set of definitions.

2

See the Applications Information section for more information about the worst other specifications for the AD6653.

3

Crosstalk is measured at 100 MHz with −1 dBFS on one channel and with no input on the alternate channel.

1

2

3

Temperature

Full 95 95 dB

AD6653BCPZ-125 AD6653BCPZ-150

Min Typ Max Min Typ Max

Unit

Rev. 0 | Page 6 of 80

AD6653

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DIGITAL SPECIFICATIONS

AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, maximum sample rate, VIN = −1.0 dBFS differential input, 1.0 V internal reference, DCS enabled, unless otherwise noted.

Table 3.

AD6653BCPZ-125 AD6653BCPZ-150

Parameter Temp

DIFFERENTIAL CLOCK INPUTS

(CLK+, CLK−) Logic Compliance CMOS/LVDS/LVPECL CMOS/LVDS/LVPECL Internal Common-Mode Bias Full 1.2 1.2 V Differential Input Voltage Full 0.2 6 0.2 6 V p-p Input Voltage Range Full AVDD − 0.3 AVDD + 1.6 AVDD − 0.3 AVDD + 1.6 V Input Common-Mode Range Full 1.1 V AVDD 1.1 V AVDD V High Level Input Voltage Full 1.2 3.6 1.2 3.6 V Low Level Input Voltage Full 0 0.8 0 0.8 V High Level Input Current Full −10 +10 −10 +10 μA Low Level Input Current Full −10 +10 −10 +10 μA Input Capacitance Full 4 4 pF Input Resistance Full 8 10 12 8 10 12 kΩ

SYNC INPUT

Logic Compliance CMOS CMOS Internal Bias Full 1.2 1.2 V Input Voltage Range Full AVDD − 0.3 AVDD + 1.6 AVDD − 0.3 AVDD + 1.6 V High Level Input Voltage Full 1.2 3.6 1.2 3.6 V Low Level Input Voltage Full 0 0.8 0 0.8 V High Level Input Current Full −10 +10 −10 +10 μA Low Level Input Current Full −10 +10 −10 +10 μA Input Capacitance Full 4 4 pF Input Resistance Full 8 10 12 8 10 12 kΩ

LOGIC INPUT (CSB)

High Level Input Voltage Full 1.22 3.6 1.22 3.6 V Low Level Input Voltage Full 0 0.6 0 0.6 V High Level Input Current Full −10 +10 −10 +10 μA Low Level Input Current Full 40 132 40 132 μA Input Resistance Full 26 26 kΩ Input Capacitance Full 2 2 pF

LOGIC INPUT (SCLK/DFS)

High Level Input Voltage Full 1.22 3.6 1.22 3.6 V Low Level Input Voltage Full 0 0.6 0 0.6 V High Level Input Current Full −92 −135 −92 −135 μA Low Level Input Current Full −10 +10 −10 +10 μA Input Resistance Full 26 26 kΩ Input Capacitance Full 2 2 pF

LOGIC INPUTS (SDIO/DCS, SMI SDFS)

High Level Input Voltage Full 1.22 3.6 1.22 3.6 V Low Level Input Voltage Full 0 0.6 0 0.6 V High Level Input Current Full −10 +10 −10 +10 μA Low Level Input Current Full 38 128 38 128 μA Input Resistance Full 26 26 kΩ Input Capacitance Full 5 5 pF

1

2

1

Min Typ Max Min Typ Max

Unit

Rev. 0 | Page 7 of 80

AD6653

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AD6653BCPZ-125 AD6653BCPZ-150

Parameter Temp

LOGIC INPUTS (SMI SDO/OEB, SMI SCLK/PDWN)

High Level Input Voltage Full 1.22 3.6 1.22 3.6 V

Low Level Input Voltage Full 0 0.6 0 0.6 V

High Level Input Current Full −90 −134 −90 −134 μA

Low Level Input Current Full −10 +10 −10 +10 μA

Input Resistance Full 26 26 kΩ

Input Capacitance Full 5 5 pF DIGITAL OUTPUTS

CMOS Mode—DRVDD = 3.3 V

High Level Output Voltage

IOH = 50 μA Full 3.29 3.29 V IOH = 0.5 mA Full 3.25 3.25 V

Low Level Output Voltage

IOL = 1.6 mA Full 0.2 0.2 V IOL = 50 μA Full 0.05 0.05 V

CMOS Mode—DRVDD = 1.8 V

High Level Output Voltage

IOH = 50 μA Full 1.79 1.79 V IOH = 0.5 mA Full 1.75 1.75 V

Low Level Output Voltage

IOL = 1.6 mA Full 0.2 0.2 V IOL = 50 μA Full 0.05 0.05 V

LVDS Mode—DRVDD = 1.8 V

Differential Output Voltage (VOD),

ANSI Mode Output Offset Voltage (VOS), ANSI Mode Full 1.15 1.25 1.35 1.15 1.25 1.35 V Differential Output Voltage (VOD),

Reduced Swing Mode Output Offset Voltage (VOS),

Reduced Swing Mode

1

Pull up.

2

Pull down.

2

Full 250 350 450 250 350 450 mV

Full 150 200 280 150 200 280 mV

Full 1.15 1.25 1.35 1.15 1.25 1.35 V

Min Typ Max Min Typ Max

Unit

Rev. 0 | Page 8 of 80

AD6653

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SWITCHING SPECIFICATIONS

Table 4.

AD6653BCPZ-125 AD6653BCPZ-150

Parameter Temperature

CLOCK INPUT PARAMETERS

Input Clock Rate Full 625 625 MHz Conversion Rate1

DCS Enabled Full 20 125 20 150 MSPS

DCS Disabled Full 10 125 10 150 MSPS CLK Period—Divide-by-1 Mode (t CLK Pulse Width High (t

Divide-by-1 Mode, DCS Enabled Full 2.4 4 5.6 2.0 3.33 4.66 ns

Divide-by-1 Mode, DCS Disabled Full 3.6 4 4.4 3.0 3.33 3.66 ns

Divide-by-2 Mode, DCS Enabled Full 1.6 1.6 ns

Divide-by-3 Through Divide-by-8 Modes,

DCS Enabled

DATA OUTPUT PARAMETERS (DATA, FD)

CMOS Noninterleaved Mode—DRVDD = 1.8 V

Data Propagation Delay (tPD)2 Full 1.6 3.9 6.2 1.6 3.9 6.2 ns

DCO Propagation Delay (t

Setup Time (tS) Full 9.5 8.16 ns

Hold Time (tH) Full 6.5 5.16 ns CMOS Noninterleaved Mode—DRVDD = 3.3 V

Data Propagation Delay (tPD)2 Full 1.9 4.1 6.4 1.9 4.1 6.4 ns

DCO Propagation Delay (t

Setup Time (tS) Full 9.7 8.36 ns

Hold Time (tH) Full 6.3 4.96 ns CMOS Interleaved and IQ Mode—DRVDD = 1.8 V

Data Propagation Delay (tPD)2 Full 1.6 3.9 6.2 1.6 3.9 6.2 ns

DCO Propagation Delay (t

Setup Time (tS) Full 4.9 4.23 ns

Hold Time (tH) Full 3.1 2.43 ns CMOS Interleaved and IQ Mode—DRVDD = 3.3 V

Data Propagation Delay (tPD) 2 Full 1.9 4.1 6.4 1.9 4.1 6.4 ns

DCO Propagation Delay (t

Setup Time (tS) Full 5.1 4.43 ns

Hold Time (tH) Full 2.9 2.23 ns LVDS Mode—DRVDD = 1.8 V

Data Propagation Delay (tPD)2 Full 2.5 4.8 7.0 2.5 4.8 7.0 ns

DCO Propagation Delay (t Pipeline Delay (Latency) NCO, FIR, fS/8 Mix Disabled Full 38 38 Cycles Pipeline Delay (Latency) NCO Enabled; FIR and fS/8

Mix Disabled (Complex Output Mode) Pipeline Delay (Latency) NCO, FIR Filter, and fS/8 Mix

Enabled

Aperture Delay (tA) Full 1.0 1.0 ns

Aperture Uncertainty (Jitter, tJ) Full 0.1 0.1 ps rms

Wake-Up Time3 Full 350 350 μs

OUT-OF-RANGE RECOVERY TIME Full 44 44 Cycles

1

Conversion rate is the clock rate after the divider.

2

Output propagation delay is measured from CLK 50% transition to DATA 50% transition, with a 5 pF load.

3

Wake-up time is dependent on the value of the decoupling capacitors.

)

CLKH

) Full 8 6.66 ns

CLK

Full 0.8 0.8 ns

) Full 4.0 5.4 7.3 4.0 5.4 7.3 ns

DCO

) Full 4.4 5.8 7.7 4.4 5.8 7.7 ns

DCO

) Full 3.4 4.8 6.7 3.4 4.8 6.7 ns

DCO

) Full 3.8 5.2 7.1 3.8 5.2 7.1 ns

DCO

) Full 3.7 5.3 7.3 3.7 5.3 7.3 ns

DCO

Full 38 38 Cycles

Full 109 109 Cycles

Min Typ Max Min Typ Max

Unit

Rev. 0 | Page 9 of 80

AD6653

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TIMING SPECIFICATIONS

Table 5.

Parameter Conditions Min Typ Max Unit

SYNC TIMING REQUIREMENTS

t

SYNC to the rising edge of CLK setup time 0.24 ns

SSYNC

t

SYNC to the rising edge of CLK hold time 0.4 ns

HSYNC

SPI TIMING REQUIREMENTS

tDS Setup time between the data and the rising edge of SCLK 2 ns tDH Hold time between the data and the rising edge of SCLK 2 ns t

Period of the SCLK 40 ns

CLK

tS Setup time between CSB and SCLK 2 ns tH Hold time between CSB and SCLK 2 ns t

Minimum period that SCLK should be in a logic high state 10 ns

HIGH

t

Minimum period that SCLK should be in a logic low state 10 ns

LOW

t

EN_SDIO

Time required for the SDIO pin to swit

ch from an input to an output

relative to the SCLK falling edge

t

DIS_SDIO

Time required for the SDIO pin to switch from an output to an input

elative to the SCLK rising edge

r

SPORT TIMING REQUIREMENTS

t

Delay from rising edge of CLK+ to rising edge of SMI SCLK 3.2 4.5 6.2 ns

CSSCLK

t

Delay from rising edge of SMI SCLK to SMI SDO −0.4 0 +0.4 ns

SSLKSDO

t

Delay from rising edge of SMI SCLK to SMI SDFS −0.4 0 +0.4 ns

SSCLKSDFS

Timing Diagrams

10 ns

CLK+

t

PD

FD BITS

t

CHANNEL A/B

DATA BITS

S

DECIMATED CMOS DAT A

DECIMATED

FD DATA

DECIMATED

DCOA/DCOB

CHANNEL A/B

FD BITS

CHANNEL A/B

Figure 2. Decimated Noninterleaved CMOS Mode D

CLK+

t

PD

DECIMATED

CMOS DATA

DECIMATED

FD DATA

DECIMATED

DCOA/DCOB

Figure 3. Decimated Noninterleaved CMOS Mode Data and Fast Detect Out

CHANNEL A/B

t

S

DATA BITS

FD BITS

t

DCO

CHANNEL A/B

FD BITS

t

H

CHANNEL A/B

FD BITS

DATA BIT S

CHANNEL A/B

FD BITS

CHANNEL A/B

FD BITS

DATA BIT S

ata and Fast Detect Output Timing (Fast Detect Mode Select Bits = 000)

t

DCO

CHANNEL A/B

DATA BITS

CHANNEL A/B

FD BITS

t

H

CHANNEL A/B

DATA BITS

CHANNEL A/B

FD BITS

put Timing (Fast Detect Mode Select Bits = 001 Through Fast Detect Mode Select Bits = 100)

06708-109

06708-002

Rev. 0 | Page 10 of 80

AD6653

www.BDTIC.com/ADI

CLK+

t

PD

t

DCO

OUTPUT DATA

DECIMATED

INTERLEAVED

CMOS DATA

DECIMATED

INTERLEAVED

FD DATA

DECIMATED

DCO

CLK+

DECIMAT ED

CMOS IQ

CMOS FD

DATA

DECIMAT ED

DCOA/DCOB

CLK–

CHANNEL A:

DATA

CHANNEL A:

FD BITS

CHANNEL A/B:

Q DATA

CHANNEL A/B:

FD BITS

CHANNEL B:

DATA

CHANNEL B:

FD BITS

Figure 4. Decimated Interleaved CMO

t

PD

CHANNEL A/B:

I DATA

CHANNEL A/B:

FD BITS

t

S

t

H

CHANNEL A:

DATA

CHANNEL A:

FD BITS

t

S

CHANNEL A/B:

Q DATA

CHANNEL A/B:

FD BITS

CHANNEL B:

DATA

CHANNEL B:

FD BITS

t

H

S Mode Data and Fast Detect Output Timing

t

DCO

CHANNEL A/B:

I DATA

CHANNEL A/B:

FD BITS

Figure 5. Decimated IQ Mode CMOS Data and Fast Detect Output Timing

CHANNEL A:

DATA

CHANNEL A:

FD BITS

CHANNEL A/B:

Q DATA

CHANNEL A/B:

FD BITS

CHANNEL B:

DATA

CHANNEL B:

FD BITS

CHANNEL A/ B:

I DATA

CHANNEL A/ B:

FD BITS

06708-003

06708-004

CLK+

LVDS

DATA

LVDS

FAST DET

DCO–

DCO+

t

PD

CHANNEL A:

DATA

CHANNEL A:

FD

Figure 6. Decimated Interleaved

CHANNEL B:

DATA

CHANNEL B:

FD

CHANNEL A:

DATA

CHANNEL A:

FD

t

DCO

CHANNEL B:

DATA

CHANNEL B:

FD

LVDS Mode Data and Fast Detect Output Timing

CHANNEL A:

DATA

CHANNEL A:

FD

06708-005

CLK+

SYNC

t

SSYNC

t

HSYNC

06708-006

Figure 7. SYNC Timing Inputs

Rev. 0 | Page 11 of 80

AD6653

S

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CLK+

CLK–

MI SCLK

SMI SDFS

SMI SDO

t

CSSCLK

t

SSCLKSDFS

Figure 8. Signal Monitor SPORT

t

SSCLKSDFS

Output Timing

DATA DATA

06708-007

Rev. 0 | Page 12 of 80

AD6653

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ABSOLUTE MAXIMUM RATINGS

Table 6.

Parameter Rating

ELECTRICAL

AVDD, DVDD to AGND −0.3 V to +2.0 V DRVDD to DRGND −0.3 V to +3.9 V AGND to DRGND −0.3 V to +0.3 V VIN+A/VIN+B, VIN−A/VIN−B to AGND −0.3 V to AVDD + 0.2 V CLK+, CLK− to AGND −0.3 V to +3.9 V SYNC to AGND −0.3 V to +3.9 V VREF to AGND −0.3 V to AVDD + 0.2 V SENSE to AGND −0.3 V to AVDD + 0.2 V CML to AGND −0.3 V to AVDD + 0.2 V RBIAS to AGND −0.3 V to AVDD + 0.2 V CSB to AGND −0.3 V to +3.9 V SCLK/DFS to DRGND −0.3 V to +3.9 V SDIO/DCS to DRGND −0.3 V to DRVDD + 0.3 V SMI SDO/OEB to DRGND −0.3 V to DRVDD + 0.3 V SMI SCLK/PDWN to DRGND −0.3 V to DRVDD + 0.3 V SMI SDFS to DRGND −0.3 V to DRVDD + 0.3 V D0A/D0B through D11A/D11B

to DRGND FD0A/FD0B through FD3A/FD3B

to DRGND DCOA/DCOB to DRGND −0.3 V to DRVDD + 0.3 V

ENVIRONMENTAL

Operating Temperature Range

(Ambient)

Maximum Junction Temperature

Under Bias

Storage Temperature Range

(Ambient)

−0.3 V to DRVDD + 0.3 V

−40°C to +85°C

150°C

−65°C to +125°C

Stresses above those listed under Absolute Maximum Ratings

y cause permanent damage to the device. This is a stress

ma rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

THERMAL CHARACTERISTICS

The exposed paddle must be soldered to the ground plane for the LFCSP package. Soldering the exposed paddle to the customer board increases the reliability of the solder joints and maximizes the thermal capability of the package.

Table 7. Thermal Resistance

Airflow Package Typ e

64-Lead LFCSP

9 mm × 9 mm (CP-64-3)

1

Per JEDEC 51-7, plus JEDEC 25-5 2S2P test board.

2

Per JEDEC JESD51-2 (still air) or JEDEC JESD51-6 (moving air).

3

Per MIL-Std 883, Method 1012.1.

4

Per JEDEC JESD51-8 (still air).

Ve

lo city

(m/s) θ

0 18.8 0.6 6.0 °C/W

1.0 16.5 °C/W

2.0 15.8 °C/W

1, 2

JA

1, 3

θ

JC

1, 4

θ

Unit

JB

Typical θJA is specified for a 4-layer PCB with a solid ground plane. As shown, airflow increases heat dissipation, which reduces θ

. In addition, metal in direct contact with the

JA

package leads from metal traces, through holes, ground, and power planes, reduces the θ

.

JA

ESD CAUTION

Rev. 0 | Page 13 of 80

AD6653

www.BDTIC.com/ADI

PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS

DRGND

D3B

D2B

D1B

D0B (LSB)

DNC

DVDD

FD3B

FD2B

FD1B

FD0B

SYNC

CSB

CLK–

CLK+

49

48

SCLK/DFS

47

SDIO/DCS

46

AVDD

45

AVDD

44

VIN+B

43

VIN–B

42

RBIAS

41

CML

40

SENSE

39

VREF

38

VIN–A

37

VIN+A

36

AVDD

35

SMI SDFS

34

SMI SCLK/PDWN

33

SMI SDO/OEB

DRVDD

D4B D5B D6B D7B D8B D9B

D10B

D11B (MSB)

DCOB DCOA

DNC DNC

D0A (LSB)

D1A D2A

646362616059585756555453525150

PIN 1

1 2 3 4 5 6 7 8

9 10 11 12 13 14 15 16

INDICAT OR

EXPOSED PADDLE, PIN 0 (BOTTO M OF PACKAGE)

AD6653

PARALLEL CMO S

TOP VIEW

(Not to Scale)

DNC = DO NOT CONNECT

171819202122232425262728293031

D3A

D4A

D5A

D6A

D7A

D8A

D9A

D10A

FD0A

DRGND

DVDD

DRVDD

FD1A

D11A (MSB)

32

FD2A

FD3A

06708-008

Figure 9. LFCSP Parallel CMOS Pin Configuration (Top View)

Table 8. Pin Function Descriptions (Parallel CMOS Mode)

Pin No. Mnemonic Type Description

ADC Power Supplies 20, 64 DRGND Ground Digital Output Ground. 1, 21 DRVDD Supply Digital Output Driver Supply (1.8 V to 3.3 V). 24, 57 DVDD Supply Digital Power Supply (1.8 V Nominal). 36, 45, 46 AVDD Supply Analog Power Supply (1.8 V Nominal). 0 AGND Ground Analog Ground. Pin 0 is the exposed thermal pad on the bottom of the package. 12, 13, 58, 59 DNC Do Not Connect. ADC Analog 37 VIN+A Input Differential Analog Input Pin (+) for Channel A. 38 VIN−A Input Differential Analog Input Pin (−) for Channel A. 44 VIN+B Input Differential Analog Input Pin (+) for Channel B. 43 VIN−B Input Differential Analog Input Pin (−) for Channel B. 39 VREF Input/Output Voltage Reference Input/Output. 40 SENSE Input Voltage Reference Mode Select. See Table 11 for details. 42 RBIAS Input/Output External Reference Bias Resistor. 41 CML Output Common-Mode Level Bias Output for Analog Inputs. 49 CLK+ Input ADC Clock Input—True. 50 CLK− Input ADC Clock Input—Complement. ADC Fast Detect Outputs 29 FD0A Output Channel A Fast Detect Indicator. See Ta ble 17 for details. 30 FD1A Output Channel A Fast Detect Indicator. See Ta ble 17 for details. 31 FD2A Output Channel A Fast Detect Indicator. See Ta ble 17 for details. 32 FD3A Output Channel A Fast Detect Indicator. See Ta ble 17 for details. 53 FD0B Output Channel B Fast Detect Indicator. See Table 17 for details. 54 FD1B Output Channel B Fast Detect Indicator. See Table 17 for details. 55 FD2B Output Channel B Fast Detect Indicator. See Table 17 for details. 56 FD3B Output Channel B Fast Detect Indicator. See Table 17 for details.

Rev. 0 | Page 14 of 80

AD6653

www.BDTIC.com/ADI

Pin No. Mnemonic Type Description

Digital Inputs 52 SYNC Input Digital Synchronization Pin. Slave mode only. Digital Outputs 14 D0A (LSB) Output Channel A CMOS Output Data. 15 D1A Output Channel A CMOS Output Data. 16 D2A Output Channel A CMOS Output Data. 17 D3A Output Channel A CMOS Output Data. 18 D4A Output Channel A CMOS Output Data. 19 D5A Output Channel A CMOS Output Data. 22 D6A Output Channel A CMOS Output Data. 23 D7A Output Channel A CMOS Output Data. 25 D8A Output Channel A CMOS Output Data. 26 D9A Output Channel A CMOS Output Data. 27 D10A Output Channel A CMOS Output Data. 28 D11A (MSB) Output Channel A CMOS Output Data. 60 D0B (LSB) Output Channel B CMOS Output Data. 61 D1B Output Channel B CMOS Output Data. 62 D2B Output Channel B CMOS Output Data. 63 D3B Output Channel B CMOS Output Data. 2 D4B Output Channel B CMOS Output Data. 3 D5B Output Channel B CMOS Output Data. 4 D6B Output Channel B CMOS Output Data. 5 D7B Output Channel B CMOS Output Data. 6 D8B Output Channel B CMOS Output Data. 7 D9B Output Channel B CMOS Output Data. 8 D10B Output Channel B CMOS Output Data. 9 D11B (MSB) Output Channel B CMOS Output Data. 11 DCOA Output Channel A Data Clock Output. 10 DCOB Output Channel B Data Clock Output. SPI Control 48 SCLK/DFS Input SPI Serial Clock/Data Format Select Pin in External Pin Mode. 47 SDIO/DCS Input/Output SPI Serial Data I/O/Duty Cycle Stabilizer Pin in External Pin Mode. 51 CSB Input SPI Chip Select. Active low. Signal Monitor Port 33 SMI SDO/OEB Input/Output Signal Monitor Serial Data Output/Output Enable Input (Active Low) in External Pin Mode.

35 SMI SDFS Output Signal Monitor Serial Data Frame Sync. 34 SMI SCLK/PDWN Input/Output Signal Monitor Serial Clock Output/Power-Down Input in External Pin Mode.

Rev. 0 | Page 15 of 80

AD6653

www.BDTIC.com/ADI

DRGND

DNC

FD3+

FD3–

FD2+

FD2–

DVDD

FD1+

FD1–

FD0+

FD0–

SYNC

CSB

CLK–

CLK+

49

48

SCLK/DFS

47

SDIO/DCS

46

AVDD

45

AVDD

44

VIN+B

43

VIN–B

42

RBIAS

41

CML

40

SENSE

39

VREF

38

VIN–A

37

VIN+A

36

AVDD

35

SMI SDFS

34

SMI SCLK/PDWN

33

SMI SDO/OEB

DRVDD

DNC

DNC D0– (LSB) D0+ (LSB )

D1– D1+ D2– D2+

DCO–

DCO+

D3– D3+ D4– D4+ D5–

646362616059585756555453525150

PIN 1

1 2 3 4 5 6 7 8

9 10 11 12 13 14 15 16

INDICATO R

EXPOSED PADDLE, PIN 0 (BOTTOM OF PACKAGE)

AD6653

PARALLEL L VDS

TOP VIEW

(Not to Scale)

DNC = DO NOT CONNECT

171819202122232425262728293031

D6–

D7–

D8–

D5+

D6+

DRGND

DRVDD

D7+

DVDD

D8+

D9–

D9+

D10–

D10+

32

D11– (MSB)

D11+ (MSB)

06708-009

Figure 10. LFCSP Interleaved Parallel LVDS Pin Configuration (Top View)

Table 9. Pin Function Descriptions (Interleaved Parallel LVDS Mode)

Pin No. Mnemonic Type Description

ADC Power Supplies 20, 64 DRGND Ground Digital Output Ground. 1, 21 DRVDD Supply Digital Output Driver Supply (1.8 V to 3.3 V). 24, 57 DVDD Supply Digital Power Supply (1.8 V Nominal). 36, 45, 46 AVDD Supply Analog Power Supply (1.8 V Nominal). 0 AGND Ground Analog Ground. Pin 0 is the exposed thermal pad on the bottom of the package. 2, 3, 62, 63 DNC Do Not Connect. ADC Analog 37 VIN+A Input Differential Analog Input Pin (+) for Channel A. 38 VIN−A Input Differential Analog Input Pin (−) for Channel A. 44 VIN+B Input Differential Analog Input Pin (+) for Channel B. 43 VIN−B Input Differential Analog Input Pin (−) for Channel B. 39 VREF Input/Output Voltage Reference Input/Output. 40 SENSE Input Voltage Reference Mode Select. See Table 1 1 for details. 42 RBIAS Input/Output External Reference Bias Resistor. 41 CML Output Common-Mode Level Bias Output for Analog Inputs. 49 CLK+ Input ADC Clock Input—True. 50 CLK− Input ADC Clock Input—Complement. ADC Fast Detect Outputs 54 FD0+ Output Channel A/Channel B LVDS Fast Detect Indicator 0—True. See Table 17 for details. 53 FD0− Output

Channel A/Channel B LVDS Fast Detect Indicator 0—Complement. See Table 1 7

r details.

fo 56 FD1+ Output Channel A/Channel B LVDS Fast Detect Indicator 1—True. See Table 17 for details. 55 FD1− Output

Channel A/Channel B LVDS Fast Detect Indicator 1—Complement. See Table 1 7

r details.

fo 59 FD2+ Output Channel A/Channel B LVDS Fast Detect Indicator 2—True. See Table 17 for details. 58 FD2− Output

Channel A/Channel B LVDS Fast Detect Indicator 2—Complement. See Table 1 7

r details.

fo 61 FD3+ Output Channel A/Channel B LVDS Fast Detect Indicator 3—True. See Table 17 for details. 60 FD3− Output

Channel A/Channel B LVDS Fast Detect Indicator 3—Complement. See Table 1 7

r details.

fo

Rev. 0 | Page 16 of 80

AD6653

www.BDTIC.com/ADI

Pin No. Mnemonic Type Description

Digital Inputs 52 SYNC Input Digital Synchronization Pin. Slave mode only. Digital Outputs 5 D0+ (LSB) Output Channel A/Channel B LVDS Output Data 0—True. 4 D0− (LSB) Output Channel A/Channel B LVDS Output Data 0—Complement. 7 D1+ Output Channel A/Channel B LVDS Output Data 1—True. 6 D1− Output Channel A/Channel B LVDS Output Data 1—Complement. 9 D2+ Output Channel A/Channel B LVDS Output Data 2—True. 8 D2− Output Channel A/Channel B LVDS Output Data 2—Complement. 13 D3+ Output Channel A/Channel B LVDS Output Data 3—True. 12 D3− Output Channel A/Channel B LVDS Output Data 3—Complement. 15 D4+ Output Channel A/Channel B LVDS Output Data 4 —True. 14 D4− Output Channel A/Channel B LVDS Output Data 4—Complement. 17 D5+ Output Channel A/Channel B LVDS Output Data 5—True. 16 D5− Output Channel A/Channel B LVDS Output Data 5—Complement. 19 D6+ Output Channel A/Channel B LVDS Output Data 6—True. 18 D6− Output Channel A/Channel B LVDS Output Data 6—Complement. 23 D7+ Output Channel A/Channel B LVDS Output Data 7—True. 22 D7− Output Channel A/Channel B LVDS Output Data 7—Complement. 26 D8+ Output Channel A/Channel B LVDS Output Data 8—True. 25 D8− Output Channel A/Channel B LVDS Output Data 8—Complement. 28 D9+ Output Channel A/Channel B LVDS Output Data 9—True. 27 D9− Output Channel A/Channel B LVDS Output Data 9—Complement. 30 D10+ Output Channel A/Channel B LVDS Output Data 10—True. 29 D10− Output Channel A/Channel B LVDS Output Data 10—Complement. 32 D11+ (MSB) Output Channel A/Channel B LVDS Output Data 11—True. 31 D11− (MSB) Output Channel A/Channel B LVDS Output Data 11—Complement. 11 DCO+ Output Channel A/Channel B LVDS Data Clock Output—True. 10 DCO− Output Channel A/Channel B LVDS Data Clock Output—Complement. SPI Control

48 SCLK/DFS Input SPI Serial Clock/Data Format Select Pin in External Pin Mode. 47 SDIO/DCS Input/Output SPI Serial Data Input/Output/Duty Cycle Stabilizer in External Pin Mode. 51 CSB Input SPI Chip Select. Active low.

Signal Monitor Port

33 SMI SDO/OEB Input/Output

35 SMI SDFS Output Signal Monitor Serial Data Frame Sync. 34 SMI SCLK/PDWN Input/Output

Signal Monitor Serial Data O Pin Mode.

Signal Monitor Serial Clock Output/Power-D Pin Mode.

utput/Output Enable Input (Active Low) in External

own Input (Active High) in External

Rev. 0 | Page 17 of 80

AD6653

V

C

S

A

V

www.BDTIC.com/ADI

EQUIVALENT CIRCUITS

LK+

IN

06708-010

Figure 11. Equivalent Analog Input Circuit

AVDD

1.2V

10kΩ 10kΩ

Figure 12. Equivalent Clock lnp

DRVDD

ut Circuit

CLK–

SCLK/DFS

Figure 15. Equivalent SCLK

SENSE

06708-011

1kΩ

26kΩ

/DFS Input Circuit

1kΩ

06708-014

06708-015

Figure 16. Equivalent SENSE Circuit

DRGND

6708-012

Figure 13. Equivalent Digital Output Circuit

DRVDD

26kΩ

DIO/DCS

1kΩ

06708-013

Figure 14. Equivalent SDIO/DCS Circuit or SMI SDFS Circuit

VDD

26kΩ

CSB

1kΩ

06708-016

Figure 17. Equivalent CSB Input Circuit

AVDD

REF

6kΩ

06708-017

Figure 18. Equivalent VREF Circuit

Rev. 0 | Page 18 of 80

AD6653

www.BDTIC.com/ADI

TYPICAL PERFORMANCE CHARACTERISTICS

AVDD = 1.8 V, DVDD = 1.8 V, DRVDD = 1.8 V, sample rate = 150 MSPS, DCS enabled, 1.0 V internal reference, 2 V p-p differential input, VIN = −1.0 dBFS, 64k sample, T the location of the second and third harmonics is noted when they fall in the pass band of the filter.

0

–20

–40

–60

–80

AMPLITUDE ( dBFS)

–100

SECOND HARMONIC

THIRD HARMONIC

= 25°C, NCO enabled, FIR filter enabled, unless otherwise noted. In the FFT plots that follow,

A

150MSPS

2.4MHz @ –1dBF S SNR = 70.9dBc (71. 9dBFS) SFDR = 83.2dBc

f

= 18.75MHz

NCO

0

150MSPS

140.1MHz @ –1dBF S SNR = 70.6dBc (71.6dBFS)

–20

SFDR = 82.9dBc

f

= 126MHz

NCO

–40

–60

–80

AMPLITUDE ( dBFS)

–100

THIRD HARMONIC

SECOND HARMONIC

–120

–140

03

FREQUENCY (MHz)

Figure 19. AD6653-150 Single-Tone FFT with f

0

150MSPS

30.3MHz @ –1dBF S SNR = 71.0dBc (72.0dBFS)

–20

SFDR = 92.3dBc

f

= 24MHz

NCO

–40

–60

–80

AMPLITUDE ( dBFS)

–100

–120

–140

03

FREQUENCY (MHz)

Figure 20. AD6653-150 Single-Tone FFT with f

0

150MSPS

70.1MHz @ –1dBF S SNR = 70.8dBc (71.8dBFS)

–20

SFDR = 82.9dBc

f

= 56MHz

NCO

–40

= 2.4 MHz, f

IN

= 30.3 MHz, f

IN

–120

5

30252015105

06708-018

= 18.75 MHz

NCO

5

30252015105

06708-019

= 24 MHz

NCO

–140

Figure 22. AD6653-150 Single-Tone FFT with f

–20

–40

–60

–80

AMPLITUDE ( dBFS)

–100

–120

–140

Figure 23. AD6653-150 Single-Tone FFT with f

–20

–40

035

0

150MSPS

220.1MHz @ –1dBF S SNR = 70.0dBc (71.0dBFS) SFDR = 80.9dBc

f

= 205MHz

NCO

03530252015105

0

150MSPS

332.1MHz @ –1dBF S SNR = 69.4dBc (70.4dBFS) SFDR = 91.2dBc

f

= 321.5MHz

NCO

FREQUENCY (MHz)

IN

THIRD HARMONIC

FREQUENCY (MHz)

IN

30252015105

= 140.1 MHz, f

= 220.1 MHz, f

= 126 MHz

NCO

= 205 MHz

NCO

06708-021

06708-022

–60

–80

AMPLITUDE ( dBFS)

–100

–120

–140

03

THIRD HARMONIC

FREQUENCY (MHz)

Figure 21. AD6653-150 Single-Tone FFT with f

= 70.1 MHz, f

IN

5

30252015105

06708-020

= 56 MHz

NCO

–60

–80

AMPLITUDE ( dBFS)

–100

–120

–140

03530252015105

Figure 24. AD6653-150 Single-Tone FFT with f

Rev. 0 | Page 19 of 80

FREQUENCY (MHz)

f

= 321.5 MHz

NCO

= 332.1 MHz,

IN

06708-023

AD6653

www.BDTIC.com/ADI

0

150MSPS

445.1MHz @ –1dBF S SNR = 69.1dBc (70.1dBFS)

–20

SFDR = 73.7dBc

f

= 429MHz

NCO

–40

–60

–80

AMPLITUDE ( dBFS)

100

–

SECOND HARMONIC

THIRD HARMONIC

0

–20

–40

–60

–80

AMPLITUDE ( dBFS)

–100

125MSPS

70.3MHz @ –1dBFS SNR = 70.9dBc (71. 9dBFS) SFDR = 85.9dBc

f

= 78MHz

NCO

THIRD HARMONIC

–120

–140

03

FREQUENCY (MHz)

Figure 25. AD6653-150 Single-Tone FFT with f

0

–20

–40

–60

–80

AMPLITUDE ( dBFS)

–100

–120

–140

SECOND HARMONIC

THIRD HARMONIC

03

FREQUENCY (MHz)

Figure 26. AD6653-125 Single-Tone FFT with f

0

125MSPS

30.3MHz @ –1dBF S SNR = 70.9dBc (71.9dBFS)

–20

SFDR = 90.7dBc

f

= 21MHz

NCO

–40

30252015105

= 445.1 MHz, f

IN

125MSPS

2.4MHz @ –1dBF S SNR = 71.0dBc (72. 0dBFS) SFDR = 84.6dBc

f

= 15.75MHz

NCO

= 2.4 MHz, f

IN

252015105

NCO

= 429 MHz

NCO

= 15.75 MHz

–120

–140

5

06708-024

030252015105

Figure 28. AD6653-125 Single-Tone FFT with f

0

–20

–40

–60

–80

AMPLITUDE ( dBFS)

–100

–120

–140

0

06708-025

030252015105

Figure 29. AD6653-125 Single-Tone FFT with f

0

–20

–40

FREQUENCY (MHz)

IN

125MSPS

140.1MHz @ –1dBF S SNR = 70.6dBc (71. 6dBFS) SFDR = 86.1dBc

f

NCO

FREQUENCY (MHz)

= 140.1 MHz, f

IN

125MSPS

220.1MHz @ –1dBF S SNR = 70.2dBc (71. 2dBFS) SFDR = 87.9dBc

f

NCO

= 70.3 MHz, f

= 142MHz

THIRD HARMONIC

= 231MHz

= 78 MHz

NCO

= 142 MHz

NCO

06708-027

06708-028

–60

–80

AMPLITUDE ( dBFS)

–100

–120

–140

03

FREQUENCY (MHz)

Figure 27. AD6653-125 Single-Tone FFT with f

252015105

= 30.3 MHz, f

IN

THIRD

HARMONIC

0

06708-026

= 21 MHz

NCO

–60

–80

AMPLITUDE ( dBFS)

–100

–120

–140

030252015105

Figure 30. AD6653-125 Single-Tone FFT with f

Rev. 0 | Page 20 of 80

FREQUENCY (MHz)

= 220.1 MHz, f

IN

= 231 MHz

NCO

06708-029

AD6653

–

www.BDTIC.com/ADI

120

95

SNR/SFDR (dBc AND dBFS)

100

80

60

40

20

0

–90 0–10–20–30–40–50–60–70–80

SNR (dBFS)

SFDR (dBc)

SFDR (dBFS)

SNR (dBc)

INPUT AMPLITUDE (dBFS)

Figure 31. AD6653-150 Single-Tone S

= 2.4 MHz, f

f

IN

120

SFDR (dBFS)

SNR (dBc)

INPUT AMPLITUDE (dBFS)

SNR/SFDR (dBc AND dBFS)

100

80

60

40

20

0

–90 0–10–20–30–40–50–60–70–80

SNR (dBFS)

SFDR (dBc)

Figure 32. AD6653-150 Single-Tone S

= 98.12 MHz, f

f

IN

85dB REFERENCE LI NE

NR/SFDR vs. Input Amplitude (A

= 18.75 MHz

NCO

85dB REFERENCE LI NE

NR/SFDR vs. Input Amplitude (A

= 100.49 MHz

NCO

) with

IN

) with

IN

90

85

80

75

SNR/SFDR (dBc)

70

65

60

0440035030025020015010050

06708-030

SFDR = –40°C

Figure 34. AD6653-125 Single-Tone SNR/SFDR vs. Input Frequency (f

SFDR = +85°C

SNR = +25°C SNR = +85°C SNR = –40°C

INPUT FREQ UENCY (MHz)

SFDR = +25°C

50

06708-033

) and

IN

Temperature with DRVDD = 3.3 V

1.5

–2.0

OFFSET

–2.5

–3.0

GAIN ERROR (%F SR)

–3.5

–4.0

–40 806040200–20

06708-031

GAIN

TEMPERATURE (° C)

0.5

0.4

0.3

0.2

0.1

0

OFFSET ERROR (%FSR)

06708-034

Figure 35. AD6653-150 Gain and Offset vs. Temperature

95

90

SFDR = +85°C

85

80

75

SNR/SFDR (dBc)

70

65

60

04

SFDR = –40°C

SNR = +25°C SNR = +85°C SNR = –40°C

INPUT FREQ UENCY (MHz)

25020015010050

SFDR = +25°C

Figure 33. AD6653-125 Single-Tone SNR/SFDR vs. Input Frequency (f

Temperature with DRVDD = 1.8 V

50400350300

06708-032

) and

IN

Rev. 0 | Page 21 of 80

0

–20

SFDR (dBc)

–40

–60

–80

SFDR/IMD3 (dBc AND dBFS )

–100

–120

–90 –78 –66 –54 –42 –30 –18 –6

IMD3 (d Bc)

SFDR (dBFS)

IMD3 (dBFS)

INPUT AMPLITUDE (dBFS )

Figure 36. AD6653-150 Two-Tone SFDR/IMD3 vs. Input Amplitude (A

= 29.12 MHz, f

f

IN1

= 32.12 MHz, fS = 150 MSPS, f

IN2

= 22 MHz

NCO

) with

IN

06708-035

AD6653

www.BDTIC.com/ADI

0

–20

SFDR (dBc)

–40

IMD3 (d Bc)

–60

–80

SFDR/IMD3 (dBc AND dBFS )

SFDR (dBFS)

–100

IMD3 (dBFS)

0

–20

–40

–60

–80

AMPLITUDE ( dBFS)

–100

–120

150MSPS

169.12MHz @ –7dBF S

172.12MHz @ –7dBF S SFDR = 83.6dBc (90.6dBFS )

f

= 177MHz

NCO

–120

–90 –78 –66 –54 –42 –30 –18 –6

INPUT AMPLITUDE (dBFS )

Figure 37. AD6653-150 Two-Tone SFDR/IMD3 vs. Input Amplitude (A

= 169.12 MHz, f

f

IN1

0

–20

–40

–60

–80

AMPLITUDE ( dBFS)

–100

–120

–140

0 5 10 15 20 25 30

Figure 38. AD6653-125, Two 64k WCDMA Carriers with f

0

150MSPS

29.12MHz @ –7dBF S

32.12MHz @ –7dBF S

–20

SFDR = 91.1dBc (98.1dBFS)

f

NCO

–40

= 172.12 MHz, fS = 150 MSPS, f

IN2

FREQUENCY ( MHz)

= 122.88 MHz, f

f

S

= 22MHz

= 168.96 MHz

NCO

= 177 MHz

NCO

= 170 MHz,

IN

) with

IN

–140

0330252015105

06708-036

Figure 40. AD6653-150 Two-Tone FFT with f

= 172.12 MHz, fS = 150 MSPS, f

f

IN2

0

–20

–40

–60

–80

AMPLITUDE ( dBFS)

–100

06708-037

–120

0 37.530.022.515.07. 5

FREQUENCY (MHz)

FREQUENCY (MHz )

= 169.12 MHz,

IN1

= 177 MHz

NCO

NPR = 61.9dBc NOTCH @ 18.5MHz NOTCH WIDT H = 3MHz

5

06708-039

06708-040

Figure 41. AD6653-150 Noise Power Ratio (NPR)

95

90

85

SFDR

–60

–80

AMPLITUDE ( dBFS)

–100

–120

–140

03

FREQUENCY (MHz)

Figure 39. AD6653-150 Two-Tone FFT with f

= 150 MSPS, f

f

S

NCO

= 29.12 MHz, f

IN1

= 22 MHz

30252015105

5

06708-038

= 32.12 MHz,

IN2

80

75

SNR/SFDR (dBc)

70

65

60

0 150125100755025

SAMPLE RATE (MSPS)

Figure 42. AD6653-150 Single-Tone SNR/SFDR vs. Sample Rate (f

f

Rev. 0 | Page 22 of 80

= 2.3 MHz

IN

SNR

) with

S

06708-041

AD6653

www.BDTIC.com/ADI

12

0.21 LSB rms

90

10

8

6

4

NUMBER OF HIT S (1M)

2

0

N – 3 N – 2 N – 1 N N + 1 N + 2 N + 3

OUTPUT CODE

Figure 43. AD6653 Grounded Input Histogram

90

85

SFDR DCS ON

80

SFDR DCS OFF

SNR DCS OFF

SNR/SFDR (dBc)

75

SNR DCS ON

70

85

SFDR

80

75

SNR/SFDR (dBc)

SNR

70

65

0.2 0.4 0. 6 0. 8 1. 0 1.2 1.4 1.6

06708-042

INPUT COMMON-MODE VOLTAGE (V)

06708-044

Figure 45. AD6653-150 SNR/SFDR vs. Input Common Mode (VCM) with

= 30.3 MHz, f

f

IN

= 45 MHz

NCO

65

20 30 40 50 60 70 80

DUTY CYCLE (%)

Figure 44. AD6653-150 SNR/SFDR vs. Duty Cycle with f

= 45 MHz

f

NCO

= 30.3 MHz,

IN

06708-043

Rev. 0 | Page 23 of 80

AD6653

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THEORY OF OPERATION

The AD6653 has two analog input channels, two decimating channels, and two digital output channels. The intermediate frequency (IF) input signal passes through several stages before appearing at the output port(s) as a filtered, decimated digital signal.

The dual ADC design can be used for diversity reception of signals,

here the ADCs operate identically on the same carrier but from

w two separate antennae. The ADCs can also be operated with independent analog inputs. The user can sample any f

/2 frequency

S

segment from dc to 150 MHz, using appropriate low-pass or band-pass filtering at the ADC inputs with little loss in ADC performance. Operation to 450 MHz analog input is permitted but occurs at the expense of increased ADC noise and distortion.

In nondiversity applications, the AD6653 can be used as a b

aseband receiver, where one ADC is used for I input data,

and the other is used for Q input data.

Synchronization capability is provided to allow synchronized timin

g between multiple channels or multiple devices. The NCO phase can be set to produce a known offset relative to another channel or device.

Programming and control of the AD6653 are accomplished usin

g a 3-bit SPI-compatible serial interface.

ADC ARCHITECTURE

AD6653 architecture consists of a front-end sample-and-hold amplifier (SHA), followed by a pipelined switched-capacitor ADC. The quantized outputs from each stage are combined into a final 12-bit result in the digital correction logic. The pipelined architecture permits the first stage to operate on a new input sample and the remaining stages to operate on the preceding samples. Sampling occurs on the rising edge of the clock.

Each stage of the pipeline, excluding the last, consists of a low

esolution flash ADC connected to a switched-capacitor digital-

r to-analog converter (DAC) and an interstage residue amplifier (MDAC). The residue amplifier magnifies the difference between the reconstructed DAC output and the flash input for the next stage in the pipeline. One bit of redundancy is used in each stage to facilitate digital correction of flash errors. The last stage simply consists of a flash ADC.

The input stage of each channel contains a differential SHA that ca

n be ac- or dc-coupled in differential or single-ended modes. The output staging block aligns the data, corrects errors, and passes the data to the output buffers. The output buffers are powered from a separate supply, allowing adjustment of the output voltage swing. During power-down, the output buffers go into a high impedance state.

ANALOG INPUT CONSIDERATIONS

The analog input to the AD6653 is a differential switchedcapacitor SHA that has been designed for optimum performance while processing a differential input signal.

The clock signal alternatively switches the SHA between sample mode and hold mode (see Figure 46). When the SHA is switched in

to sample mode, the signal source must be capable of charging

the sample capacitors and settling within 1/2 of a clock cycle.

A small resistor in series with each input can help reduce the

eak transient current required from the output stage of the

p driving source. A shunt capacitor can be placed across the inputs to provide dynamic charging currents. This passive network creates a low-pass filter at the ADC input; therefore, the precise values are dependent on the application.

In IF undersampling applications, any shunt capacitors should b

e reduced. In combination with the driving source impedance, the shunt capacitors limit input bandwidth. Refer to Application Note AN-742, Frequency Domain Response of Switched-

Capacitor ADCs; Application Note AN-827, A Resonant Approach to Interfacing Amplifiers to Switched-Capacitor ADCs; and the Analog Dialogue article, “

Wi

deband A/D Converters,” for more information on this subject

(see www.analog.com).

Transformer-Coupled Front-End for

In general, the precise values are

dependent on the application.

S

C

H

C

H

S

06708-048

VIN+

VIN–

C

PIN, PAR

C

PIN, PAR

Figure 46. Switched-Capac

S

C

S

H

C

S

itor SHA Input

For best dynamic performance, the source impedances driving VIN+ and VIN− should be matched such that common-mode settling errors are symmetrical. These errors are reduced by the common-mode rejection of the ADC.

An internal differential reference buffer creates positive and n

egative reference voltages that define the input span of the ADC core. The output common mode of the reference buffer is set to VCMREF (approximately 1.6 V).

Input Common Mode

The analog inputs of the AD6653 are not internally dc biased. In ac-coupled applications, the user must provide this bias externally. Setting the device so that V

= 0.55 × AVDD is

CM

recommended for optimum performance, but the device functions over a wider range with reasonable performance (see Figure 45). An o

n-board common-mode voltage reference is included in the design and is available from the CML pin. Optimum performance is achieved when the common-mode voltage of the analog input is set by the CML pin voltage (typically 0.55 × AVDD).

Rev. 0 | Page 24 of 80

AD6653

Ω

F

V

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Differential Input Configurations

Optimum performance is achieved while driving the AD6653 in a differential input configuration. For baseband applications, the

AD8138, ADA4937-2, and ADA4938-2 differential drivers provide

exce

llent performance and a flexible interface to the ADC. The

output common-mode voltage of the

e CML pin of the AD6653 (see Figure 47), and the driver can

th be co

nfigured in a Sallen-Key filter topology to provide band

AD8138 is easily set with

limiting of the input signal.

499

1V p-p

0.1µF

49.9Ω 499Ω

AD8138

523Ω

499Ω

Figure 47. Differential Input Configuration Using the AD8138

R

C

R

VIN+

AD6653

VIN–

AVDD

CML

For baseband applications where SNR is a key parameter, differential transformer coupling is the recommended input configuration. An example is shown in

nalog input, the CML voltage can be connected to the center

a

Figure 48. To bias the

tap of the secondary winding of the transformer.

R

2V p-p

49.9Ω

0.1µF

C

R

Figure 48. Differential Transformer-Coupled Configuration

VIN+

AD6653

VIN–

CML

06708-050

2V p-p

0.1µ

S

SP

A

Figure 49. Differential Double Balun Input Configuration

0.1µ

P

0.1µF

CC

06708-049

The signal characteristics must be considered when selecting

nsformer. Most RF transformers saturate at frequencies

a tra below a few megahertz (MHz). Excessive signal power can also cause core saturation, which leads to distortion.

At input frequencies in the second Nyquist zone and above, the

oise performance of most amplifiers is not adequate to achieve

n the true SNR performance of the AD6653. For applications where SNR is a key parameter, differential double balun coupling is the recommended input configuration (see

Figure 49).

An alternative to using a transformer coupled input at frequencies

he second Nyquist zone is to use the AD8352 differential

in t dr

iver, as shown in Figure 50. See the AD8352 data sheet for

m

ore information. In addition, if the application requires an amplifier with variable gain, the AD8375 or AD8376 digital va

riable gain amplifiers (DVGAs) provide good performance

driving the AD6653.

In any configuration, the value of the shunt capacitor, C, is dep

endent on the input frequency and source impedance and may need to be reduced or removed. Ta ble 10 displays r

ecommended values to set the RC network. However, these values are dependent on the input signal and should be used only as a starting guide.

Table 10. Example RC Network

R Series

Frequency Range (MHz)

(Ω Each) C Differential (pF)

0 to 70 33 15 70 to 200 33 5 200 to 300 15 5 >300 15 Open

25Ω

0.1µF

R

C

R

VIN+

AD6653

VIN–

CML

06708-051

ANALOG INPUT

0.1µF

0Ω

16

1

2

R

C

D

0.1µF

R

D

G

3

4

5

0Ω

AD8352

14

8, 13

10

0.1µF

11

0.1µF

200Ω

0.1µF

R

C

R

0.1µF

Figure 50. Differential Input Configuration Using the AD8352

Rev. 0 | Page 25 of 80

VIN+

AD6653

VIN–

CML

06708-052

AD6653

A

V

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Single-Ended Input Configuration

A single-ended input can provide adequate performance in cost-sensitive applications. In this configuration, SFDR and distortion performance degrade due to the large input commonmode swing. If the source impedances on each input are matched, there should be little effect on SNR performance. a ty

pical single-ended input configuration.

2V p-p

10µF

49.9Ω

10µF

0.1µF

Figure 51. Single-Ended Input Configuration

AVD D

1kΩ

DD

R

C

R

Figure 51 shows

VIN+

AD6653

VIN–

06708-053

VOLTAGE REFERENCE

A stable and accurate voltage reference is built into the AD6653. The input range can be adjusted by varying the reference voltage applied to the AD6653, using either the internal reference or an externally applied reference voltage. The input span of the ADC tracks reference voltage changes linearly. The various reference modes are summarized in the sections that follow. The Deco

upling section describes the best practices PCB layout of

t

he reference.

Internal Reference Connection

A comparator within the AD6653 detects the potential at the SENSE pin and configures the reference into four possible modes, which are summarized in Tabl e 1 1 . If SENSE is grounded, the re

ference amplifier switch is connected to the internal resistor

divider (see

ENSE pin to VREF switches the reference amplifier output to

S

Figure 52), setting VREF to 1.0 V. Connecting the

the SENSE pin, completing the loop and providing a 0.5 V reference output.

Reference

If a resistor divider is connected externally to the chip, as shown in Figure 53, the switch again sets to the SENSE pin. This puts t

he reference amplifier in a noninverting mode with the VREF

output defined as follows:

The input range of the ADC always equals twice the voltage at

he reference pin for either an internal or an external reference.

t

VIN+A/ VIN+B

VIN–A/VIN–B

ADC

CORE

VREF

0.1µF1. 0µF

SENSE

Figure 52. Internal Reference Configuration

R2

⎛

+×=

.VREF 150

⎜

R1

⎝

VIN+A/VIN+ B

VIN–A/ VIN–B

0.1µF1. 0µF

R2

SENSE

R1

⎞ ⎟ ⎠

VREF

SELECT

LOGIC

AD6653

SELECT

LOGIC

0.5V

ADC

CORE

6708-054

Figure 53. Programmable Reference Configuration

Table 11. Reference Configuration Summary

Selected Mode SENSE Voltage Resulting VREF (V) Resulting Differential Span (V p-p)

External Reference AVDD N/A 2 × external reference

Internal Fixed Reference VREF 0.5 1.0

R2

⎞

⎛

Programmable Reference 0.2 V to VREF ⎟

15.0

⎜ ⎝

+×

R1

⎠

(see Figure 53)

2 × VREF

Internal Fixed Reference AGND to 0.2 V 1.0 2.0

Rev. 0 | Page 26 of 80

AD6653

6708-055

AD6653

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If the internal reference of the AD6653 is used to drive multiple converters to improve gain matching, the loading of the reference by the other converters must be considered. Figure 54 depicts

ow the internal reference voltage is affected by loading.

h

0

VREF = 0.5V

–0.25

VREF = 1.0V

–0.50

CLOCK INPUT CONSIDERATIONS

For optimum performance, the AD6653 sample clock inputs, CLK+ and CLK−, should be clocked with a differential signal. The signal is typically ac-coupled into the CLK+ and CLK− pins via a transformer or capacitors. These pins are biased internally (see Figure 56) and require no external bias.

AVDD

1.2V

CLK–CLK+

–0.75

–1.00

REFERENCE VOL TAGE ERROR ( %)

–1.25

02

0.5 1.0 1.5

LOAD CURRENT (mA)

.0

6708-056

Figure 54. VREF Accuracy vs. Load

External Reference Operation

The use of an external reference may be necessary to enhance the gain accuracy of the ADC or improve thermal drift characteristics. Figure 55 shows the typical drift characteristics of the

ternal reference in both 1.0 V and 0.5 V modes.

in

2.5

2.0

1.5

1.0

0.5

0

–0.5

–1.0

–1.5

REFERENCE VOL TAGE ERROR ( mV)

–2.0

–2.5

–40

–200 20406080

TEMPERATURE (° C)

06708-057

Figure 55. Typical VREF Drift

When the SENSE pin is tied to AVDD, the internal reference is disabled, allowing the use of an external reference. An internal reference buffer loads the external reference with an equivalent 6 kΩ load (see Figure 18). The internal buffer generates the p

ositive and negative full-scale references for the ADC core. Therefore, the external reference must be limited to a maximum of 1.0 V.

2pF 2pF

6708-058

Figure 56. Equivalent Clock Input Circuit

Clock Input Options

The AD6653 has a very flexible clock input structure. Clock input can be a CMOS, LVDS, LVPECL, or sine wave signal. Regardless of the type of signal being used, the clock source jitter is of the most concern, as described in the

Jitter Considerations section.

Figure 57 and Figure 58 show two preferred methods for clocking th

e AD6653 (at clock rates up to 625 MHz). A low jitter clock source is converted from a single-ended signal to a differential signal, using an RF transformer. The back-to-back Schottky diodes across the transformer secondary limit clock excursions into the AD6653 to approximately 0.8 V p-p differential. This helps prevent the large voltage swings of the clock from feeding through to other portions of the AD6653 while preserving the fast rise and fall times of the signal, which are critical to low jitter performance.

XFMR

0.1µF

®

0.1µF0.1µF

0.1µF

0.1µF1nF

0.1µF

SCHOTTKY

DIODES:

HSMS2822

SCHOTTKY

DIODES:

HSMS2822

CLK+

AD6653

CLK–

CLK+

ADC

AD6653

CLK–

ADC

06708-157

Mini-Circuits

ADT1–1WT, 1:1Z

CLOCK

INPUT

50Ω

100Ω

Figure 57. Transformer-Coupled Differential Clock (Up to 200 MHz)

CLOCK

INPUT

50Ω

1nF

Figure 58. Balun-Coupled Differential Clock (Up to 625 MHz)

06708-059

Rev. 0 | Page 27 of 80

AD6653

[

]

×−=

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If a low jitter clock source is not available, another option is to ac-couple a differential PECL signal to the sample clock input pins as shown in Figure 59. The AD9510/AD9511/AD9512/

AD9513/AD9514/AD9515/AD9516 clock drivers offer excellent

jitter performance.

CLOCK

INPUT

CLOCK

INPUT

50kΩ 50kΩ

0.1µF

AD951x

PECL DRIVER

0.1µF

Figure 59. Differential PECL Sample Clock (Up to 625 MHz)

0.1µF CLK+

100Ω

0.1µF

240Ω240Ω

ADC

AD6653

CLK–

A third option is to ac-couple a differential LVDS signal to the sample clock input pins, as shown in Figure 60. The AD9510/

AD9511/AD9512/AD9513/AD9514/AD9515/AD9516 clock

drivers offer excellent jitter performance.

CLOCK

INPUT

CLOCK

INPUT

50kΩ 50kΩ

0.1µF

AD951x

LVDS DRIVER

0.1µF

Figure 60. Differential LVDS Sample Clock (Up to 625 MHz)

0.1µF

100Ω

0.1µF

CLK+

ADC

AD6653

CLK–

In some applications, it may be acceptable to drive the sample clock inputs with a single-ended CMOS signal. In such applications, the CLK+ pin should be driven directly from a CMOS gate, and the CLK− pin should be bypassed to ground with a 0.1 F capacitor in parallel with a 39 kΩ resistor (see Figure 61). CLK+ can be driven directly from a CMOS gate. Although the CLK+ input circuit supply is AVDD (1.8 V), this input is designed to withstand input voltages of up to 3.6 V, making the selection of the drive logic voltage very flexible.

V

CC

0.1µF

1kΩ

CMOS DRIVE R

1kΩ

AD951x

CMOS DRIVE R

AD951x

CLOCK

INPUT

CLOCK

INPUT

50Ω

Figure 61. Single-Ended 1.8 V CMOS Sample Clock (Up to 150 MSPS)

V

CC

0.1µF

50Ω

Figure 62. Single-Ended 3.3 V CMOS Sample Clock (Up to 150 MSPS)

0.1µF

OPTIONAL

100Ω

OPTIO NAL

100Ω

39kΩ

0.1µF

CLK+

ADC

AD6653

CLK–

CLK+

ADC

AD6653

CLK–

06708-060

06708-061

06708-062

06708-063

Input Clock Divider

The AD6653 contains an input clock divider with the ability to divide the input clock by integer values between 1 and 8. If a divide ratio other than 1 is selected, the duty cycle stabilizer is automatically enabled.

The AD6653 clock divider can be synchronized using the external SYNC input. Bit 1 and Bit 2 of Register 0x100 allow the clock divider to be resynchronized on every SYNC signal or only on the first SYNC signal after the register is written. A valid SYNC causes the clock divider to reset to its initial state.

This synchronization feature allows multiple parts to have their clock dividers aligned to guarantee simultaneous input sampling.

Clock Duty Cycle

Typical high speed ADCs use both clock edges to generate a variety of internal timing signals and, as a result, may be sensitive to clock duty cycle. Commonly, a ±5% tolerance is required on the clock duty cycle to maintain dynamic performance characteristics.

The AD6653 contains a duty cycle stabilizer (DCS) that retimes the nonsampling (falling) edge, providing an internal clock signal with a nominal 50% duty cycle. This allows the user to provide a wide range of clock input duty cycles without affecting performance of the AD6653. Noise and distortion performance are nearly flat for a wide range of duty cycles with the DCS on, as shown in Figure 44.

Jitter on the rising edge of the input clock is still of paramount concern and is not easily reduced by the internal stabilization circuit. The duty cycle control loop does not function for clock rates less than 20 MHz nominally. The loop has a time constant associated with it that must be considered when the clock rate can change dynamically. A wait time of 1.5 µs to 5 µs is required after a dynamic clock frequency increase or decrease before the DCS loop is relocked to the input signal. During the time period that the loop is not locked, the DCS loop is bypassed, and internal device timing is dependent on the duty cycle of the input clock signal. In such applications, it may be appropriate to disable the duty cycle stabilizer. In all other applications, enabling the DCS circuit is recommended to maximize ac performance.

Jitter Considerations

High speed, high resolution ADCs are sensitive to the quality of the clock input. The degradation in SNR at a given input frequency (f

) due to jitter (tJ) can be calculated by

IN

tπfSNR

2log20

JIN

In the equation, the rms aperture jitter represents the root mean square of all jitter sources, which include the clock input, the analog input signal, and the ADC aperture jitter specification. IF undersampling applications are particularly sensitive to jitter, as shown in Figure 63.

Rev. 0 | Page 28 of 80

AD6653

www.BDTIC.com/ADI

75

70

MEASURED

65

60

SNR (dBc)

55

50

45

1 10 100 1000

INPUT FREQUENCY (MHz)

0.05ps

0.20ps

0.50ps

1.00ps

1.50ps

2.00ps

2.50ps

3.00ps

06708-064

Figure 63. SNR vs. Input Frequency and Jitter

The clock input should be treated as an analog signal in cases where aperture jitter may affect the dynamic range of the AD6653. Power supplies for clock drivers should be separated from the ADC output driver supplies to avoid modulating the clock signal with digital noise. Low jitter, crystal-controlled oscillators make the best clock sources. If the clock is generated from another type of source (by gating, dividing, or another method), it should be retimed by the original clock at the last step.

Refer to Application Note AN-50

1 and Application Note AN-756 for more information about jitter performance as it relates to ADCs (see

www.analog.com).

POWER DISSIPATION AND STANDBY MODE

As shown in Figure 64 and Figure 65, the power dissipated by the AD6653 is proportional to its sample rate. In CMOS output mode, the digital power dissipation is determined primarily by the strength of the digital drivers and the load on each output bit. The maximum DRVDD current (I

I

DRVDD

where N is t

= V

× f

DRVDD

CLK

× N

he number of output bits (26, in the case of the

AD6653, assuming the FD bits are inactive).

This maximum current occurs when every output bit switches on e

very clock cycle, that is, a full-scale square wave at the Nyquist

frequency of f

/2. In practice, the DRVDD current is established

CLK

by the average number of output bits switching, which is determined by the sample rate and the characteristics of the analog input signal. Reducing the capacitive load presented to the output drivers can minimize digital power consumption. The data in Figure 64 and Figure 65 was taken using the same operating co

nditions as those used for the Typi c a l Pe r formance

aracteristics, with a 5 pF load on each output driver.

Ch

) can be calculated by

DRVDD

1.50

1.25

I

AVDD

1.00

0.75

0.50

TOTAL POWER (W)

0.25

0

0 50 100

SAMPLE RATE (MSPS)

TOTAL POWER

I

DVDD

I

DRVDD

0.6

0.5

0.4

0.3

0.2

SUPPLY CURRENT (A)

0.1

0

15025 75 125

Figure 64. AD6653-150 Power and Current vs. Sample Rate

1.50

1.25

1.00 I

AVDD

0.75

0.50

TOTAL POWER (W)

0.25

0

050100

TOTAL POWER

I

DVDD

I

DRVDD

SAMPLE RATE (MSPS)

0.6

0.5

0.4

0.3

0.2

SUPPLY CURRENT (A)

0.1

0

12525 75

Figure 65. AD6653-125 Power and Current vs. Sample Rate

By asserting PDWN (either through the SPI port or by asserting the PDWN pin high), the AD6653 is placed in power-down mode. In this state, the ADC typically dissipates 2.5 mW. During power-down, the output drivers are placed in a high impedance state. Asserting the PDWN pin low returns the AD6653 to its normal operating mode. Note that PDWN is referenced to the digital output driver supply (DRVDD) and should not exceed that supply voltage level. PDWN can be driven with 1.8 V logic, even when DRVDD is at 3.3 V.

Low power dissipation in power-down mode is achieved by shutting down the reference, reference buffer, biasing networks, and clock. Internal capacitors are discharged when entering power-down mode and then must be recharged when returning to normal operation. As a result, the wake-up time is related to the time spent in power-down mode, and shorter power-down cycles result in proportionally shorter wake-up times.

06708-065

06708-066

Rev. 0 | Page 29 of 80

AD6653

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When using the SPI port interface, the user can place the ADC in power-down mode or standby mode. Standby mode allows the user to keep the internal reference circuitry powered when faster wake-up times are required. See the Memory Map Register Description section and Application Note AN-877, Interfacing to High Speed ADCs via SPI at www.analog.com for additional details.

DIGITAL OUTPUTS

The AD6653 output drivers can be configured to interface with

1.8 V to 3.3 V CMOS logic families by matching DRVDD to the digital supply of the interfaced logic. Alternatively, the AD6653 outputs can be configured for either ANSI LVDS or reduced drive LVDS using a 1.8 V DRVDD supply.

In CMOS output mode, the output drivers are sized to provide sufficient output current to drive a wide variety of logic families. However, large drive currents tend to cause current glitches on the supplies that may affect converter performance. Applications requiring the ADC to drive large capacitive loads or large fanouts may require external buffers or latches.

The output data format can be selected for either offset binary or twos complement by setting the SCLK/DFS pin when operating in the external pin mode (see Table 12). As detailed in Application Note AN-877, Interfacing to High Speed ADCs via SPI, the data format can be selected for offset binary, twos complement, or gray code when using the SPI control.

Table 12. SCLK/DFS Mode Selection (External Pin Mode)

Voltage at Pin SCLK/DFS SDIO/DCS

AGND (default) Offset binary DCS disabled AVDD Twos complement DCS enabled

Digital Output Enable Function (OEB)

The AD6653 has a flexible, three-state ability for the digital output pins. The three-state modeis enabled using the SMI SDO/OEB pin or through the SPI interface.

If the SMI SDO/OEB pin is low, the output data drivers are enabled. If the SMI SDO/OEB pin is high, the output data drivers are placed in a high impedance state.

This OEB function is not intended for rapid access to the data bus. Note that OEB is referenced to the digital output driver supply (DRVDD) and should not exceed that supply voltage.

When using the SPI interface, the data and fast detect outputs of each channel can be independently three-stated by using the output enable bar bit, Bit 4 in Register 0x14.

Interleaved CMOS Mode

Setting Bit 5 in Register 0x14 enables interleaved CMOS output mode. In this mode, output data is routed through Port A with the ADC Channel A output data present on the rising edge of DCO and the ADC Channel B output data present on the falling edge of DCO.

Timing

The AD6653 provides latched data with a pipeline delay that is dependent on which of the digital back end features are enabled. Data outputs are available one propagation delay (t rising edge of the clock signal.

The length of the output data lines and loads placed on them should be minimized to reduce transients within the AD6653. These transients can degrade converter dynamic performance.

The lowest typical conversion rate of the AD6653 is 10 MSPS. At clock rates below 10 MSPS, dynamic performance may degrade.

) after the

PD

Data Clock Output (DCO)

The AD6653 also provides data clock output (DCO) intended for capturing the data in an external register. Figure 2 through Figure 6 show a graphical timing description of the AD6653 output modes.

Table 13. Output Data Format

Input (V) Condition (V) Offset Binary Output Mode Twos Complement Mode OR

VIN+ – VIN− < −VREF − 0.5 LSB 0000 0000 0000 1000 0000 0000 1 VIN+ – VIN− = −VREF 0000 0000 0000 1000 0000 0000 0 VIN+ – VIN− = 0 1000 0000 0000 0000 0000 0000 0 VIN+ – VIN− = +VREF − 1.0 LSB 1111 1111 1111 0111 1111 1111 0 VIN+ − VIN− > +VREF − 0.5 LSB 1111 1111 1111 0111 1111 1111 1

Rev. 0 | Page 30 of 80

AD6653

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DIGITAL DOWNCONVERTER

The AD6653 includes a digital processing section that provides filtering and reduces the output data rate. This digital processing section includes a numerically controlled oscillator (NCO), a half-band decimating filter, an FIR filter, and a second coarse NCO (f

/8 fixed value) for output frequency translation. Each

ADC

of these processing blocks (except the decimating half-band filter) has control lines that allow it to be independently enabled and disabled to provide the desired processing function. The digital downconverter can be configured to output either real data or complex output data. These blocks can be configured in five recommended combinations to implement different signal processing functions.

DOWNCONVERTER MODES

Tabl e 1 4 details the recommended downconverter modes of operation in the AD6653.

Table 14. Downconverter Modes

Mode NCO/Filter Output Type

1 Half-band filter only Real 2 Half-band filter and FIR filter Real 3 NCO and half-band filter Complex 4 NCO, half-band filter, and FIR filter Complex 5

NCO, half-band filter, FIR filter, and f

/8 NCO

ADC

Real

NUMERICALLY CONTROLLED OSCILLATOR (NCO)

Frequency translation is accomplished with an NCO. Each of the two processing channels shares a common NCO. Amplitude and phase dither can be enabled on chip to improve the noise and spurious performance of the NCO. A phase offset word is available to create a known phase relationship between multiple AD6653s.

Because the decimation filter prevents usage of half the Nyquist s

pectrum, a means is needed to translate the sampled input spectrum into the usable range of the decimation filter. To achieve this, a 32-bit, fine tuning, complex NCO is provided. This NCO/mixer allows the input spectrum to be tuned to dc, where it can be effectively filtered by the subsequent filter blocks to prevent aliasing.

a maximum usable bandwidth of 16.5 MHz when using the filter in real mode (NCO bypassed) or a maximum usable bandwidth of 33.0 MHz when using the filter in the complex mode (NCO enabled).

The optional fixed-coefficient FIR filter provides additional f

iltering capability to sharpen the half-band roll-off to enhance the alias protection. It removes the negative frequency images to avoid aliasing negative frequencies for real outputs.

f

/8 FIXED-FREQUENCY NCO

ADC

A fixed f signal from dc to f

/8 NCO is provided to translate the filtered, decimated

ADC

/8 to allow a real output. Figure 66 to

ADC

Figure 69 show an example of a 20 MHz input as it is processed

y the blocks of the AD6653.

b

–50 –24 –14 14 24 50–4 40

Figure 66. Example AD6653 Real 20 MHz Bandwidth Input Signal Centered at

14 M

Hz (f

= 100 MHz)

ADC

06708-067

–50 –38 –28 0 10 50–18 –10

Figure 67. Example AD6653 20 MHz Bandwidth Input Signal Tuned to

DC Using the

NCO (NCO Frequency = 14 MHz)

06708-068

–50 –38 –28 0 10 50–18 –10

Figure 68. Example AD6653 20 MHz Bandwidth Input Signal with the

Negativ

e Image Filtered by the Half-Band and FIR Filters

06708-069

HALF-BAND DECIMATING FILTER AND FIR FILTER

The goal of the AD6653 digital filter block is to allow the sample rate to be reduced by a factor of 2 while rejecting aliases that fall into the band of interest. The half-band filter is designed to operate as either a low-pass or high-pass filter and to provide greater

–50 0.25 22.512.5 50

Figure 69. Example AD6653 20 MHz Bandwidth Input Signal Tuned to

than 100 dB of alias protection for 22% of the input rate of the structure. For an ADC sample rate of 150 MSPS, this provides

Rev. 0 | Page 31 of 80

/8 for Real Output

f

ADC

6708-070

AD6653

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NUMERICALLY CONTROLLED OSCILLATOR (NCO)

FREQUENCY TRANSLATION

This processing stage comprises a digital tuner consisting of a 32-bit complex numerically controlled oscillator (NCO). The two channels of the AD6653 share a single NCO. The NCO is optional and can be bypassed by clearing Bit 0 of Register 0x11D. This NCO block accepts a real input from the ADC stage and outputs a frequency translated complex (I and Q) output.

The NCO frequency is programmed in Register 0x11E,

ister 0x11F, Register 0x120, and Register 0x121. These four

Reg 8-bit registers make up a 32-bit unsigned frequency programming word. Frequencies between −CLK/2 and +CLK/2 are represented using the following frequency words:

• 0x8000 0000 represents a frequency given by −CLK/2.

• 0x0000 0000 represents dc (frequency = 0 Hz).

• 0x7FFF FFFF represents CLK/2 − CLK/2

Use the following equation to calculate the NCO frequency:

f

CLK

ffMod

CLK

32

NCO_FREQ

where:

NC

O_FREQ is a 32-bit twos complement number representing

the NCO frequency register. f is the desired carrier frequency in hertz (Hz).

is the AD6653 ADC clock rate in hertz (Hz).

f

CLK

×=

2

32

.

),(

NCO SYNCHRONIZATION

The AD6653 NCOs within a single part or across multiple parts can be synchronized using the external SYNC input. Bit 3 and Bit 4 of Register 0x100 allow the NCO to be resynchronized on every SYNC signal or only on the first SYNC signal after the register is written. A valid SYNC causes the NCO to restart at the programmed phase offset value.

PHASE OFFSET

The NCO phase offset register at Address 0x122 and Address 0x123 adds a programmable offset to the phase accumulator of the NCO. This 16-bit register is interpreted as a 16-bit unsigned integer. A 0x00 in this register corresponds to no offset, and a 0xFFFF corresponds to an offset of 359.995°. Each bit represents a phase change of 0.005°. This register allows multiple NCOs to be synchronized to produce outputs with predictable phase differences. Use the following equation to calculate the NCO phase offset value:

NCO_PHASE = 2

where:

O_PHASE is a decimal number equal to the 16-bit binary

NC

number to be programmed at Register 0x122 and Register 0x123. PHASE is the desired NCO phase in degrees.

16

× PHASE/360

NCO AMPLITUDE AND PHASE DITHER

The NCO block contains amplitude and phase dither to improve the spurious performance. Amplitude dither improves performance by randomizing the amplitude quantization errors within the angular-to-Cartesian conversion of the NCO. This option reduces spurs at the expense of a slightly raised noise floor. With amplitude dither enabled, the NCO has an SNR of >93 dB and an SFDR of >115 dB. With amplitude dither disabled, the SNR is increased to >96 dB at the cost of SFDR performance, which is reduced to 100 dB. The NCO amplitude dither is recommended and is enabled by setting Bit 1 of Register 0x11D.

Rev. 0 | Page 32 of 80

AD6653

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DECIMATING HALF-BAND FILTER AND FIR FILTER

The goal of the AD6653 half-band digital filter is to allow the sample rate to be reduced by a factor of 2 while rejecting aliases that fall into the band of interest. This filter is designed to operate as either a low-pass or a high-pass filter and to provide >100 dB of alias protection for 11% of the input rate of the structure. Used in conjunction with the NCO and the FIR filter, the halfband filter can provide an effective band-pass. For an ADC sample rate of 150 MSPS, this provides a maximum usable bandwidth of 33 MHz.

HALF-BAND FILTER COEFFICIENTS

The 19-tap, symmetrical, fixed-coefficient half-band filter has low power consumption due to its polyphase implementation. Table 15 lists the coefficients of the half-band filter. The normalized coefficients used in the implementation and the decimal equivalent value of the coefficients are also listed. Coefficients not listed in Table 15 are 0s.

Table 15. Fixed Coefficients for Half-Band Filter

Coefficient Number

Normalized Coefficient

Decimal Coefficient (20-Bit)

C0, C18 0.0008049 844 C2, C16 −0.0059023 −6189 C4, C14 0.0239182 25080 C6, C12 −0.0755024 −79170 C8, C10 0.3066864 321584 C9 0.5 524287

HALF-BAND FILTER FEATURES

In the AD6653, the half-band filter cannot be disabled. The filter can be set for a low-pass or high-pass response. For a highpass filter, Bit 1 of Register 0x103 should be set; for a low-pass response, this bit should be cleared. The low-pass response of the filter with respect to the normalized output rate is shown in Figure 70, and the high-pass response is shown in Figure 71.

0 –10 –20 –30 –40 –50 –60 –70

AMPLITUDE ( d Bc)

–80 –90

–100 –110

0 0.1 0.2 0.3 0.4

FRACTION OF INPUT SAMPLE RATE

Figure 70. Half-Band Filter, Low-Pass Response

06708-071

The half-band filter has a ripple of 0.000182 dB and a rejection of 100 dB. For an alias rejection of 100 dB, the alias protected bandwidth is 11% of the input sample rate. If both the I and the Q paths are used, a complex bandwidth of 22% of the input rate is available.

In the event of even Nyquist zone sampling, the half-band filter can be configured to provide a spectral reversal. Setting Bit 2 high in Address 0x103 enables the spectral reversal feature.

The half-band decimation phase can be selected such that the half-band filter starts on the first or second sample following synchronization. This shifts the output from the half-band between the two input sample clocks. The decimation phase can be set to 0 or 1, using Bit 3 of Register 0x103.

FIXED-COEFFICIENT FIR FILTER

Following the half-band filters is a 66-tap, fixed-coefficient FIR filter. This filter is useful in providing extra alias protection for the decimating half-band filter. It is a simple sum-of-products FIR filter with 66 filter taps and 21-bit fixed coefficients. Note that this filter does not decimate. The normalized coefficients used in the implementation and the decimal equivalent value of the coefficients are listed in Table 16.

The user can either select or bypass this filter, but the FIR filter can be enabled only when the half-band filter is enabled. Writing Logic 0 to the enable FIR filter bit (Bit 0) in Register 0x102 bypasses this fixed-coefficient filter. The filter is necessary when using the final NCO with a real output; bypassing it when using other configurations results in power savings.

0 –10 –20 –30 –40 –50 –60 –70

AMPLITUDE ( d Bc)

–80 –90

–100 –110

0 0.1 0.2 0.3 0.4

FRACTION OF INPUT SAMPLE RATE

Figure 71. Half-Band Filter, High-Pass Response

06708-072

Rev. 0 | Page 33 of 80

AD6653

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Table 16. FIR Filter Coefficients

Coefficient Number

C0, C65 0.0001826 383 C1, C64 0.0006824 1431 C2, C63 0.0009298 1950 C3, C62 0.0000458 96 C4, C61 −0.0012689 −2661 C5, C60 −0.0008345 −1750 C6, C59 0.0011806 2476 C7, C58 0.0011387 2388 C8, C57 −0.0018439 −3867 C9, C56 −0.0024557 −5150 C10, C55 0.0018063 3788 C11, C54 0.0035825 7513 C12, C53 −0.0021510 −4511 C13, C52 −0.0056810 −11914 C14, C51 0.0017405 3650 C15, C50 0.0078602 16484 C16, C49 −0.0013437 −2818 C17, C48 −0.0110626 −23200 C18, C47 −0.0000229 −48 C19, C46 0.0146618 30748 C20, C45 0.0018959 3976 C21, C44 −0.0195594 −41019 C22, C43 −0.0053153 −11147 C23, C42 0.0255623 53608 C24, C41 0.0104036 21818 C25, C40 −0.0341468 −71611 C26, C39 −0.0192165 −40300 C27, C38 0.0471258 98830 C28, C37 0.0354118 74264 C29, C36 −0.0728111 −152696 C30, C35 −0.0768890 −161248 C31, C34 0.1607208 337056 C32, C33 0.4396725 922060

Normalized Coefficient

Decimal Coefficient (21-Bit)

SYNCHRONIZATION

The AD6653 half-band filters within a single part or across multiple parts can be synchronized using the external SYNC input. Bit 5 and Bit 6 of Register 0x100 allow the half-bands to be resynchronized on every SYNC signal or only on the first SYNC signal after the register is written. A valid SYNC causes the half-band filter to restart at the programmed decimation phase value.

COMBINED FILTER PERFORMANCE

The combined response of the half-band filter and the FIR filter is shown in Figure 72. The act of bandlimiting the ADC data w

ith the half-band filter ideally provides a 3 dB improvement in the SNR at the expense of the sample rate and available bandwidth of the output data. As a consequence of finite math, additional quantization noise is added to the system due to truncation in the NCO and half-band. As a consequence of the digital filter rejection of out-of-band noise (assuming no quantization in the filters and with a white noise floor from the ADC), there should be a 3.16 dB improvement in the ADC SNR. However, the added quantization lessens improvement to about 2.66 dB.

0

–10

–20

–30

–40

–50

–60

–70

AMPLITUDE ( dBc)

–80

–90

–100

–110

0 0.1 0.2 0.3 0.4

Figure 72. Half-Band Filter and FIR Filter Composite Response

FRACTION OF INPUT SAMPLE RATE

06708-073

FINAL NCO

The output of the 32-bit fine tuning NCO is complex and typically centered in frequency around dc. This complex output is carried through the stages of the half-band and FIR filters to provide proper antialiasing filtering. The final NCO provides a means to move this complex output signal away from dc so that a real output can be provided from the AD6653. The final NCO, if enabled, translates the output from dc to a frequency equal to the ADC sampling frequency divided by 8 (f the user a decimated output signal centered at f Optionally, this final NCO can be bypassed, and the dc-centered I and Q values can be output in an interleaved fashion.

/8). This provides

ADC

/8 in frequency.

ADC

Rev. 0 | Page 34 of 80

AD6653

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ADC OVERRANGE AND GAIN CONTROL

In receiver applications, it is desirable to have a mechanism to reliably determine when the converter is about to be clipped. The standard overflow indicator provides after-the-fact information on the state of the analog input that is of limited usefulness. Therefore, it is helpful to have a programmable threshold below full scale that allows time to reduce the gain before the clip actually occurs. In addition, because input signals can have significant slew rates, latency of this function is of major concern. Highly pipelined converters can have significant latency. A good compromise is to use the output bits from the first stage of the ADC for this function. Latency for these output bits is very low, and overall resolution is not highly significant. Peak input signals are typically between full scale and 6 dB to 10 dB below full scale. A 3-bit or 4-bit output provides adequate range and resolution for this function.

Using the SPI port, the user can provide a threshold above which

n overrange output is active. As long as the signal is below that

a threshold, the output should remain low. The fast detect outputs can also be programmed via the SPI port so that one of the pins functions as a traditional overrange pin for customers who currently use this feature. In this mode, all 14 bits of the converter are examined in the traditional manner, and the output is high for the condition normally defined as overflow. In either mode, the magnitude of the data is considered in the calculation of the condition (but the sign of the data is not considered). The threshold detection responds identically to positive and negative signals outside the desired range (magnitude).

FAST DETECT OVERVIEW

The AD6653 contains circuitry to facilitate fast overrange detection, allowing very flexible external gain control implementations. Each ADC has four fast detect (FD) output pins that are used to output information about the current state of the ADC input level. The function of these pins is programmable via the fast detect mode select bits and the fast detect enable bit in Register 0x104, allowing range information to be output from several points in the internal data path. These output pins can also be set up to indicate the presence of overrange or underrange conditions, according to programmable threshold levels.

hows the six configurations available for the fast detect pins.

s

Tabl e 1 7

Table 17. Fast Detect Mode Select Bit Settings

Fast Detec t Mode Select Bits (Register 0x104[3:1])

000 ADC fast magnitude (see Table 18) 001

010

011

100 OR C_UT F_UT F_LT 101 OR F_UT IG DG

1

The fast detect pins are FD0A/FD0B to FD3A/FD3B for the CMOS mode

configuration and FD0+/FD0− to FD3+/FD3− for the LVDS mode configuration.

2

See the ADC Overrange (OR) and Gain Switching sections for more

information about OR, C_UT, F_UT, F_LT, IG, and DG.

Information Presented on Fast Detect (FD) Pins of Each ADC

FD[3] FD[2] FD[1] FD[0]

ADC fast magnitude

(see Tab le 19)

ADC fast

magnitude

(see Tab le 20)

ADC fast

tude

magni

(see Tab le 20)

OR F_LT

C_UT F_LT

1, 2

OR

ADC FAST MAGNITUDE

When the fast detect output pins are configured to output the ADC fast magnitude (that is, when the fast detect mode select bits are set to 0b000), the information presented is the ADC level from an early converter stage with a latency of only two clock cycles in CMOS output modes. In LVDS output mode, the fast detect bits have a latency of six cycles in all fast detect modes. Using the fast detect output pins in this configuration provides the earliest possible level indication information. Because this information is provided early in the datapath, there is significant uncertainty in the level indicated. The nominal levels, along with the uncertainty indicated by the ADC fast magnitude, are shown in

ecause the DCO is at one-half the sample rate, the user can

B obtain the fast detect information by sampling the fast detect outputs on both the rising and falling edges of DCO (see Figure 2

r timing information).

fo

Table 18. ADC Fast Magnitude Nomimal Levels with Fast Detect Mode Select Bits = 000

ADC Fast Magitude on FD[3:0] Pins

0000 <−24 Minimum to −18.07 0001 −24 to −14.5 −30.14 to −12.04 0010 −14.5 to −10 −18.07 to −8.52 0011 −10 to −7 −12.04 to −6.02 0100 −7 to −5 −8.52 to −4.08 0101 −5 to −3.25 −6.02 to −2.5 0110 −3.25 to −1.8 −4.08 to −1.16 0111 −1.8 to −0.56 −2.5 to FS 1000 −0.56 to 0 −1.16 to 0

Nominal Input Magnitude Below FS (dB)

Nominal Input Magnitude Uncertainty (dB)

Tabl e 1 8 .

Rev. 0 | Page 35 of 80

AD6653

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When the fast detect mode select bits are set to 0b001, 0b010, or 0b011, a subset of the fast detect output pins is available. In these modes, the fast detect output pins have a latency of six clock cycles, and the greater of the two input samples is output at the DCO rate. Table 19 shows the corresponding ADC input levels when the fast detect mode select bits are set to 0b001 (that is, when the ADC fast magnitude is presented on the FD[3:1] pins).

Table 19. ADC Fast Magnitude Nomimal Levels with Fast Detect Mode Select Bits = 001

ADC Fast Magitude on FD[2:0] Pins

000 <−24 Minimum to −18.07 001 −24 to −14.5 −30.14 to −12.04 010 −14.5 to −10 −18.07 to −8.52 011 −10 to −7 −12.04 to −6.02 100 −7 to −5 −8.52 to −4.08 101 −5 to −3.25 −6.02 to −2.5 110 −3.25 to −1.8 −4.08 to −1.16 111 −1.8 to 0 −2.5 to 0

When the fast detect mode select bits are set to 0b010 or 0b011 (that is, when ADC fast magnitude is presented on the FD[2:1] pins), the LSB is not provided. The input ranges for this mode are shown in Table 20.

Table 20. ADC Fast Magnitude Nomimal Levels with Fast Detect Mode Select Bits = 010 or 011

ADC Fast Magitude on FD[2:1] Pins

00 <−14.5 Minimum to −12.04 01 −14.5 to −7 −18.07 to −6.02 10 −7 to −3.25 −8.52 to −2.5 11 −3.25 to 0 −4.08 to 0

Nominal Input Magnitude Below FS (dB)

Nominal Input Magnitude Uncertainty (dB)

ADC OVERRANGE (OR)

The ADC overrange indicator is asserted when an overrange is detected on the input of the ADC. The overrange condition is determined at the output of the ADC pipeline and, therefore, is subject to a latency of 12 ADC clock cycles. An overrange at the input is indicated by this bit 12 clock cycles after it occurs.

GAIN SWITCHING

The AD6653 includes circuitry that is useful in applications either where large dynamic ranges exist or where gain ranging converters are employed. This circuitry allows digital thresholds to be set such that an upper threshold and a lower threshold can be programmed. Fast detect mode select bits = 010 through fast detect mode select bits = 101 support various combinations of the gain switching options.

One such use is to detect when an ADC is about to reach full scale with a particular input condition. The result is to provide an indicator that can be used to quickly insert an attenuator that prevents ADC overdrive.

Coarse Upper Threshold (C_UT)

The coarse upper threshold indicator is asserted if the ADC fast magnitude input level is greater than the level programmed in the coarse upper threshold register (Address 0x105[2:0]). This value is compared with the ADC Fast Magnitude Bits[2:0]. The coarse upper threshold output is output two clock cycles after the level is exceeded at the input and, therefore, provides a fast indication of the input signal level. The coarse upper threshold levels are shown in Table 21. This indicator remains asserted for a minimum of two ADC clock cycles or until the signal drops below the threshold level.

Table 21. Coarse Upper Threshold Levels

C_UT Is Active When Signal Coarse Upper Threshold Register[2:0]

000 <−24 001 −24 010 −14.5 011 −10 100 −7 101 −5 110 −3.25 111 −1.8

Magnitude Below FS

Is Greater Than (dB)

Fine Upper Threshold (F_UT)

The fine upper threshold indicator is asserted if the input magnitude exceeds the value programmed in the fine upper threshold register located in Register 0x106 and Register 0x107. The 13-bit threshold register is compared with the signal magnitude at the output of the ADC. This comparison is subject to the ADC clock latency but is accurate in terms of converter resolution. The fine upper threshold magnitude is defined by the following equation:

13

dBFS = 20 log(Threshold Magnitude/2

)

Fine Lower Threshold (F_LT)

The fine lower threshold indicator is asserted if the input magnitude is less than the value programmed in the fine lower threshold register located at Register 0x108 and Register 0x109. The fine lower threshold register is a 13-bit register that is compared with the signal magnitude at the output of the ADC. This comparison is subject to ADC clock latency but is accurate in terms of converter resolution. The fine lower threshold magnitude is defined by the following equation:

13

dBFS = 20 log(Threshold Magnitude/2

The operation of the fine upper threshold and fine lower threshold indicators is shown in Figure 73.

)

Rev. 0 | Page 36 of 80

AD6653

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Increment Gain (IG) and Decrement Gain (DG)

The increment gain and decrement gain indicators are intended to be used together to provide information to enable external gain control. The decrement gain indicator works in conjunction with the coarse upper threshold bits, asserting when the input magnitude is greater than the 3-bit value in the coarse upper threshold register (Address 0x105). The increment gain indicator, similarly, corresponds to the fine lower threshold bits except that it is asserted only if the input magnitude is less than the value programmed in the fine lower threshold register after the dwell time elapses. The dwell time is set by the 16-bit dwell time value located at Address 0x10A and Address 0x10B and is set in units of ADC input clock cycles ranging from 1 to 65,535. The fine lower threshold register is a 13-bit register that

TIMER RESET BY

RISE ABOVE F_LT

is compared with the magnitude at the output of the ADC. This comparison is subject to the ADC clock latency but allows a finer, more accurate comparison. The fine upper threshold magnitude is defined by the following equation:

dBFS = 20 log(Thresho

ld Magnitude/2

13

)

The decrement gain output works from the ADC fast detect

pins, providing a fast indication of potential overrange

output conditions. The increment gain uses the comparison at the output of the ADC, requiring the input magnitude to remain below an accurate, programmable level for a predefined period before signaling external circuitry to increase the gain.

The operation of the increment gain output and decrement gain output

DWELL TIME

is shown graphically in

UPPER THRESHOL D (COARSE OR F INE)

Figure 73.

FINE LOW ER THRESHOL D

C_UT OR F_UT*

F_LT

DG

IG

*C_UT AND F_UT DIFF ER ONLY I N ACCURACY AND LATENCY.

NOTE: OUT PUTS FO LLOW THE INSTANT ANEOUS SIG NAL LEVEL AND NOT THE E NVELOPE BUT ARE GUARANTEED ACTIVE FOR A MINIMUM OF 2 ADC CLOCK CYCLE S.

Figure 73. Threshold Settings for C_UT

, F_UT, F_LT, DG, and IG

DWELL TIME

TIMER COMPLETES BEF ORE SIGNAL RISES ABOVE F_LT

6708-074

Rev. 0 | Page 37 of 80

AD6653

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SIGNAL MONITOR

The signal monitor block provides additional information about the signal being digitized by the ADC. The signal monitor computes the rms input magnitude, the peak magnitude, and/or the number of samples by which the magnitude exceeds a particular threshold. Together, these functions can be used to gain insight into the signal characteristics and to estimate the peak/average ratio or even the shape of the complementary cumulative distribution function (CCDF) curve of the input signal. This information can be used to drive an AGC loop to optimize the range of the ADC in the presence of real-world signals.

The signal monitor result values can be obtained from the part by

eading back internal registers at Address 0x116 to Address 0x11B,

r using the SPI port or the signal monitor SPORT output. The output contents of the SPI-accessible signal monitor registers are set via the two signal monitor mode bits of the signal monitor control register (Address 0x112). Both ADC channels must be configured for the same signal monitor mode. Separate SPI-accessible, 20-bit signal monitor result (SMR) registers are provided for each ADC channel. Any combination of the signal monitor functions can also be output to the user via the serial SPORT interface. These outputs are enabled using the peak detector output enable, the rms magnitude output enable, and the threshold crossing output enable bits in the signal monitor SPORT control register (Address 0x111).

For each signal monitor measurement, a programmable signal

itor period register (SMPR) controls the duration of the

mon measurement. This time period is programmed as the number of input clock cycles in a 24-bit signal monitor period register located at Address 0x113, Address 0x114, and Address 0x115. This register can be programmed with a period from 128 samples

24

to 16.78 (2

) million samples.

Because the dc offset of the ADC can be significantly larger t

han the signal of interest (affecting the results from the signal monitor), a dc correction circuit is included as part of the signal monitor block to null the dc offset before measuring the power.

PEAK DETECTOR MODE

The magnitude of the input port signal is monitored over a programmable time period (determined by SMPR) to give the peak value detected. This function is enabled by programming a Logic 1 in the signal monitor mode bits of the signal monitor control register or by setting the peak detector output enable bit in the signal monitor SPORT control register. The 24-bit SMPR must be programmed before activating this mode.

After enabling this mode, the value in the SMPR is loaded into a moni

tor period timer, and the countdown is started. The magnitude of the input signal is compared with the value in the internal peak level holding register (not accessible to the user), and the greater of the two is updated as the current peak level. The initial value of the peak level holding register is set to the

current ADC input signal magnitude. This comparison continues until the monitor period timer reaches a count of 1.

When the monitor period timer reaches a count of 1, the 13-bit peak

level value is transferred to the signal monitor holding register (not accessible to the user), which can be read through the SPI port or output through the SPORT serial interface. The monitor period timer is reloaded with the value in the SMPR, and the countdown is restarted. In addition, the magnitude of the first input sample is updated in the peak level holding register, and the comparison and update procedure, as explained previously, continues.

Figure 74 is a block diagram of the peak detector logic. The

MR register contains the absolute magnitude of the peak

S detected by the peak detector logic.

FROM

MEMORY

MAP

FROM

INPUT

PORTS

LOAD

CLEAR

DOWN

COUNTER

IS COUNT = 1?

POWER MONITOR

HOLDING

REGISTER

POWER MONI TOR

PERIOD REGISTER

MAGNITUDE

STORAGE REGISTER

LOAD LOAD

COMPARE

A>B

Figure 74. ADC Input Peak Detector Block Diagram

TO

INTERRUPT

CONTRO LLER

TO

MEMORY

MAP

RMS/MS MAGNITUDE MODE

In this mode, the root-mean-square (rms) or mean-square (ms) magnitude of the input port signal is integrated (by adding an accumulator) over a programmable time period (determined by SMPR) to give the rms or ms magnitude of the input signal. This mode is set by programming Logic 0 in the signal monitor mode bits of the signal monitor control register or by setting the rms magnitude output enable bit in the signal monitor SPORT control register. The 24-bit SMPR, representing the period over which integration is performed, must be programmed before activating this mode.

After enabling the rms/ms magnitude mode, the value in the S

MPR is loaded into a monitor period timer, and the countdown is started immediately. Each input sample is converted to floating-point format and squared. It is then converted to 11-bit, fixed-point format and added to the contents of the 24-bit accumulator. The integration continues until the monitor period timer reaches a count of 1.

When the monitor period timer reaches a count of 1, the square r

oot of the value in the accumulator is taken and transferred (after some formatting) to the signal monitor holding register, which can be read through the SPI port or output through the SPORT serial port. The monitor period timer is reloaded with the value in the SMPR, and the countdown is restarted.

06708-075

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AD6653

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In addition, the first input sample signal power is updated in the accumulator, and the accumulation continues with the subsequent input samples. Figure 75 illustrates the rms

agnitude monitoring logic.

m

FROM

MEMORY

MAP

POWER MONITOR

PERIOD REGISTER

FROM INPUT

PORTS

Figure 75. ADC Input RMS Magnitude Monitoring Block Diagram

ACCUMULATOR

DOWN

COUNTER

LOAD

CLEAR LOAD

IS COUNT = 1?

POWER MONITOR

HOLDING

REGISTER

TO

INTERRUPT

CONTRO LLER

TO

MEMORY

MAP

For rms magnitude mode, the value in the signal monitor result (SMR) register is a 20-bit fixed-point number. The following equation can be used to determine the rms magnitude in dBFS from the MAG value in the register. Note that if the signal monitor period (SMP) is a power of 2, the second term in the equation becomes 0.

RMS Magnitude = 20 log

SMPMAG

⎛ ⎜ ⎝

⎞

−

⎟

2

⎠

⎡

log10

[]

⎢

2

⎣

⎤

)(log20

SMPceil

⎥

2

⎦

For ms magnitude mode, the value in the SMR is a 20-bit fixedp

oint number. The following equation can be used to determine the ms magnitude in dBFS from the MAG value in the register. Note that if the SMP is a power of 2, the second term in the equation becomes 0.

MS Magnitude

= 10 log

SMPMAG

⎞

⎛ ⎜ ⎝

−

⎟

2

⎠

⎡

log10

[]

⎢

2

⎣

⎤

)(log20

SMPceil

⎥

2

⎦

THRESHOLD CROSSING MODE

In the threshold crossing mode of operation, the magnitude of the input port signal is monitored over a programmable time period (given by SMPR) to count the number of times it crosses a certain programmable threshold value. This mode is set by programming Logic 1x (where x is a don’t care bit) in the signal monitor mode bits of the signal monitor control register or by setting the threshold crossing output enable bit in the signal monitor SPORT control register. Before activating this mode, the user needs to program the 24-bit SMPR and the 13-bit upper threshold register for each individual input port. The same upper threshold register is used for both signal monitoring and gain control (see the

on).

secti

After entering this mode, the value in the SMPR is loaded into

tor period timer, and the countdown is started. The magni-

a moni tude of the input signal is compared with the upper threshold register (programmed previously) on each input clock cycle. If the input signal has a magnitude greater than the upper threshold register, the internal count register is incremented by 1. The initial value of the internal count register is set to 0. This comparison and incrementing of the internal count register continues until the monitor period timer reaches a count of 1.

ADC Overrange and Gain Control

06708-076

When the monitor period timer reaches a count of 1, the value

he internal count register is transferred to the signal monitor

in t holding register, which can be read through the SPI port or output through the SPORT serial port.

The monitor period timer is reloaded with the value in the S

MPR register, and the countdown is restarted. The internal

count register is also cleared to a value of 0. Figure 76 illustrates

he threshold crossing logic. The value in the SMR register is

t the number of samples that have a magnitude greater than the threshold register.

FROM

MEMORY

MAP

FROM INPUT

PORTS

FROM

MEMORY

MAP

B

DOWN

COUNTER

LOAD

CLEAR

COMPARE

A>B

IS COUNT = 1?

LOAD

POWER MO NITO R

HOLDING

REGISTER

POWER MONITO R

PERIOD REGISTER

A

COMPARE

A>B

UPPER

THRESHOL D

REGIST ER

Figure 76. ADC Input Threshold Crossing Block Diagram

TO

INTERRUPT

CONTROL LER

TO

MEMORY

MAP

ADDITIONAL CONTROL BITS

For additional flexibility in the signal monitoring process, two control bits are provided in the signal monitor control register. They are the signal monitor enable bit and the complex power calculation mode enable bit.

Signal Monitor Enable Bit

The signal monitor enable bit, located in Bit 0 of Register 0x112, enables operation of the signal monitor block. If the signal monitor function is not needed in a particular application, this bit should be cleared to conserve power.

Complex Power Calculation Mode Enable Bit

When this bit is set, the part assumes that Channel A is digitizing the I data and Channel B is digitizing the Q data for a complex input signal (or vice versa). In this mode, the power reported is equal to

22

QI +

This result is presented in the Signal Monitor DC Value Channel A r

egister if the signal monitor mode bits are set to 00. The Signal Monitor DC Value Channel B register continues to compute the Channel B value.

DC CORRECTION

Because the dc offset of the ADC may be significantly larger than the signal being measured, a dc correction circuit is included to null the dc offset before measuring the power. The dc correction circuit can also be switched into the main signal path, but this may not be appropriate if the ADC is digitizing a time-varying signal with significant dc content, such as GSM.

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DC Correction Bandwidth

The dc correction circuit is a high-pass filter with a programmable bandwidth (ranging between 0.15 Hz and 1.2 kHz at 125 MSPS). The bandwidth is controlled by writing the 4-bit dc correction control register located at Register 0x10C, Bits[5:2]. The following equation can be used to compute the bandwidth value for the dc correction circuit:

BWCorrDC

π×

2

f

−−

14k

CLK

×=

2__

where: k is t

he 4-bit value programmed in Bits[5:2] of Register 0x10C (values between 0 and 13 are valid for k; programming 14 or 15 provides the same result as programming 13).

is the AD6653 ADC sample rate in hertz (Hz).

f

CLK

DC Correction Readback

The current dc correction value can be read back in Register 0x10D and Register 0x10E for Channel A and Register 0x10F and Register 0x110 for Channel B. The dc correction value is a 14-bit value that can span the entire input range of the ADC.

DC Correction Freeze

Setting Bit 6 of Register 0x10C freezes the dc correction at its current state and continues to use the last updated value as the dc correction value. Clearing this bit restarts dc correction and adds the currently calculated value to the data.

DC Correction Enable Bits

Setting Bit 0 of Register 0x10C enables dc correction for use in the signal monitor calculations. The calculated dc correction value can be added to the output data signal path by setting Bit 1 of Register 0x10C.

SIGNAL MONITOR SPORT OUTPUT

The SPORT is a serial interface with three output pins: the SMI SCLK (SPORT clock), SMI SDFS (SPORT frame sync), and SMI SDO (SPORT data output). The SPORT is the master and drives all three SPORT output pins on the chip.

SMI SCLK

The data and frame sync are driven on the positive edge of the SMI SCLK. The SMI SCLK has three possible baud rates: 1/2, 1/4, or 1/8 the ADC clock rate, based on the SPORT controls. The SMI SCLK can also be gated off when not sending any data, based on the SPORT SMI SCLK sleep bit. Using this bit to disable the SMI SCLK when it is not needed can reduce any coupling errors back into the signal path, if these prove to be a problem in the system. Doing so, however, has the disadvantage of spreading the frequency content of the clock. If desired the SMI SCLK can be left running to ease frequency planning.

SMI SDFS

The SMI SDFS is the serial data frame sync, and it defines the start of a frame. One SPORT frame includes data from both datapaths. The data from Datapath A is sent just after the frame sync, followed by data from Datapath B.

SMI SDO

The SMI SDO is the serial data output of the block. The data is sent MSB first on the next positive edge after the SMI SDFS. Each data output block includes one or more of rms magnitude, peak level, and threshold crossing values from each datapath in the stated order. If enabled, the data is sent, rms first, followed by peak and threshold, as shown in

GATED, BASED O N CONTROL

Figure 77.

SMI SCLK

SMI SDFS

SMI SDO

SMI SCLK

SMI SDFS

SMI SDO

RMS/M S CH A

MSB MSB

20 CYCLES 16 CYCL ES16 CYCL ES 20 CYCLES 16 CYCLES 16 CYCLES

MSB MSBRMS/MS CH A RMS/MS CH ALSB THR CH A RMS/ MS CH B LSB THR CH B

LSB LSB

20 CYCLES 16 CYCL ES 20 CYCLES 16 CYCLES

PK CH A PK CH B

Figure 77. Signal Monitor SPORT Output Timing

Figure 78. Signal Monitor SPORT Output Ti

THR CH A

(RMS, Peak, and Threshold Enabled)

GATED, BASED ON CONTROL

ming (RMS and Threshold Enabled)

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THR CH BRMS/M S CH B

RMS/MS CH A

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CHANNEL/CHIP SYNCHRONIZATION

The AD6653 has a SYNC input that allows the user flexible

synchronization options for synchronizing the internal blocks.

The sync feature is useful for guaranteeing synchronized operation

across multiple ADCs. The input clock divider, NCO, half-band

filters, and signal monitor block can be synchronized using the

SYNC input. Each of these blocks, except for the signal monitor,

can be enabled to synchronize on a single occurrence of the

SYNC signal or on every occurrence.

The SYNC input is internally synchronized to the sample clock. However, to ensure that there is no timing uncertainty between multiple parts, the SYNC input signal should be synchronized to the input clock signal. The SYNC input should be driven using a single-ended CMOS-type signal.

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SERIAL PORT INTERFACE (SPI)

The AD6653 serial port interface (SPI) allows the user to configure the converter for specific functions or operations through a structured register space provided inside the ADC. The SPI gives the user added flexibility and customization, depending on the application. Addresses are accessed using the serial port and can be written to or read from via the port. Memory is organized into bytes that can be further divided into fields. These fields are documented in the Memory Map section. For detailed operational information, see Application Note AN-877, Interfacing to High Speed ADCs via SPI, at

www.analog.com.

CONFIGURATION USING THE SPI

Three pins define the SPI of this ADC: the SCLK/DFS pin, the SDIO/DCS pin, and the CSB pin (see Table 22). The SCLK/DFS (serial clock) pin is used to synchronize the read and write data presented from/to the ADC. The SDIO/DCS (serial data input/ output) pin is a dual-purpose pin that allows data to be sent and read from the internal ADC memory map registers. The CSB (chip select bar) pin is an active-low control that enables or disables the read and write cycles.

Table 22. Serial Port Interface Pins

Pin Function

SCLK

SDIO

CSB

The falling edge of the CSB, in conjunction with the rising edge of the SCLK, determines the start of the framing. An example of the serial timing and its definitions can be found in Figure 79 and Table 5.

Other modes involving the CSB are available. The CSB can be held low indefinitely, which permanently enables the device; this is called streaming. The CSB can stall high between bytes to allow for additional external timing. When CSB is tied high, SPI functions are placed in a high impedance mode. This mode turns on any SPI pin secondary functions.

During an instruction phase, a 16-bit instruction is transmitted. Data follows the instruction phase and its length is determined by the W0 bit and the W1 bit.

Serial Clock. The serial shift clock input, which is used to synchronize serial interface reads and writes. Serial Data Input/Output. A dual-purpose pin that typically serves as an input or an output, depending on the instruction being sent and the relative position in the timing frame. Chip Select Bar. An active-low control that gates the read and write cycles.

All data is composed of 8-bit words. The first bit of each individual byte of serial data indicates whether a read command or a write command is issued. This allows the serial data input/output (SDIO) pin to change direction from an input to an output.

In addition to word length, the instruction phase determines whether the serial frame is a read or write operation, allowing the serial port to be used both to program the chip and to read the contents of the on-chip memory. If the instruction is a readback operation, performing a readback causes the serial data input/ output (SDIO) pin to change direction from an input to an output at the appropriate point in the serial frame.

Data can be sent in MSB-first mode or in LSB-first mode. MSB first is the default on power-up and can be changed via the SPI port configuration register. For more information about this and other features, see Application Note AN-877, Interfacing to High Speed ADCs via SPI, at www.analog.com.

HARDWARE INTERFACE

The pins described in Table 22 comprise the physical interface between the user programming device and the serial port of the AD6653. The SCLK pin and the CSB pin function as inputs when using the SPI interface. The SDIO pin is bidirectional, functioning as an input during write phases and as an output during readback.

The SPI interface is flexible enough to be controlled by either FPGAs or microcontrollers. One method for SPI configuration is described in detail in Application Note AN-812, Microcontroller- Based Serial Port Interface (SPI) Boot Circuit.

The SPI port should not be active during periods when the full dynamic performance of the converter is required. Because the SCLK signal, the CSB signal, and the SDIO signal are typically asynchronous to the ADC clock, noise from these signals can degrade converter performance. If the on-board SPI bus is used for other devices, it may be necessary to provide buffers between this bus and the AD6653 to prevent these signals from transitioning at the converter inputs during critical sampling periods.

Some pins serve a dual function when the SPI interface is not being used. When the pins are strapped to AVDD or ground during device power-on, they are associated with a specific function. The Digital Outputs section describes the strappable functions supported on the AD6653.

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CONFIGURATION WITHOUT THE SPI

In applications that do not interface to the SPI control registers,

the SDIO/DCS pin, the SCLK/DFS pin, the SMI SDO/OEB pin,

and the SMI SCLK/PDWN pin serve as standalone CMOS-

compatible control pins. When the device is powered up, it is

assumed that the user intends to use the pins as static control

lines for the duty cycle stabilizer, output data format, output

enable, and power-down feature control. In this mode, the CSB

chip select should be connected to AVDD, which disables the

serial port interface.

Table 23. Mode Selection

Pin External

Voltage

AVDD (default) Duty cycle stabilizer enabled SDIO/DCS AGND Duty cycle stabilizer disabled AVDD Twos complement enabled SCLK/DFS AGND (default) Offset binary enabled AVDD Outputs in high impedance SMI SDO/OEB AGND

(default)

SMI SCLK/PDWN

AVDD

AGND (default)

Configuration

Outputs enabled

Chip in power-down or standby

Normal operation

SPI ACCESSIBLE FEATURES

Table 24 provides a brief description of the general features that are accessible via the SPI. These features are described in Application Note AN-877, Interfacing to High Speed ADCs via SPI (see www.analog.com). The AD6653 part-specific features are described in the Memory Map Register Description section.

Table 24. Features Accessible Using the SPI

Feature Name Description

Modes

Clock Allows the user to access the DCS via the SPI Offset

Tes t I /O

Output Mode Allows the user to set up outputs Output Phase Allows the user to set the output clock polarity Output Delay Allows the user to vary the DCO delay VREF Allows the user to set the reference voltage.

Allows the user to set either power-down mode or standby mode

Allows the user to digitally adjust the converter offset

Allows the user to set test modes to have known data on output bits

CSB

SCLK

SDIO

DON’T CARE

t

DS

t

S

R/W W1 W0 A12 A11 A10 A9 A8 A7

t

DH

HIGH

t

LOW

Figure 79. Serial Port Interface Timing Diagram

t

CLK

D5 D4 D3 D2 D1 D0

t

H

DON’T CARE

DON’T CAREDON’T CARE

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MEMORY MAP

READING THE MEMORY MAP REGISTER TABLE

Each row in the memory map register table has eight bit locations. The memory map is roughly divided into four sections: the chip configuration registers (Address 0x00 to Address 0x02); the channel index and transfer registers (Address 0x05 and Address 0xFF); the ADC functions registers, including setup, control, and test (Address 0x08 to Address 0x18); and the digital feature control registers (Address 0x100 to Address 0x123).

The memory map register table (see Table 25) documents the default hexadecimal value for each hexadecimal address shown. The column with the heading Bit 7 (MSB) is the start of the default hexadecimal value given. For example, Address 0x18, the VREF select register, has a hexadecimal default value of 0xC0. This means that Bit 7 = 1, Bit 6 = 1, and the remaining bits are 0s. This setting is the default reference selection setting. The default value uses a 2.0 V p-p reference. For more information on this function and others, see Application Note AN-877, Interfacing to High Speed ADCs via SPI. This document details the functions controlled by Register 0x00 to Register 0xFF. The remaining registers, from Register 0x100 to Register 0x123, are documented in the Memory Map Register Description section.

Open Locations

All address and bit locations that are not included in Table 25 are not currently supported for this device. Unused bits of a valid address location should be written with 0s. Writing to these locations is required only when part of an address location is open (for example, Address 0x18). If the entire address location is open (for example, Address 0x13), this address location should not be written.

Default Values

After the AD6653 is reset, critical registers are loaded with default values. The default values for the registers are given in the memory map register table, Table 25.

Logic Levels

An explanation of logic level terminology follows:

• “Bit is set” is synonymous with “bit is set to Logic 1” or

“writing Logic 1 for the bit.”

• “Clear a bit” is synonymous with “bit is set to Logic 0” or

“writing Logic 0 for the bit.”

Transfer Register Map

Address 0x08 to Address 0x18 and Address 0x11E to Address 0x123 are shadowed. Writes to these addresses do not affect part operation until a transfer command is issued by writing 0x01 to Address 0xFF, setting the transfer bit. This allows these registers to be updated internally and simultaneously when the transfer bit is set. The internal update takes place when the transfer bit is set, and the bit autoclears.

Channel-Specific Registers

Some channel setup functions, such as the signal monitor thresholds, can be programmed differently for each channel. In these cases, channel address locations are internally duplicated for each channel. These registers and bits are designated in Table 25 as local. These local registers and bits can be accessed by setting the appropriate Channel A or Channel B bits in Register 0x05. If both bits are set, the subsequent write affects the registers of both channels. In a read cycle, only Channel A or Channel B should be set to read one of the two registers. If both bits are set during an SPI read cycle, the part returns the value for Channel A. Registers and bits designated as global in Table 25 affect the entire part or the channel features where independent settings are not allowed between channels. The settings in Register 0x05 do not affect the global registers and bits.

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MEMORY MAP REGISTER TABLE

All address and bit locations that are not included in Tabl e 25 are not currently supported for this device.

Table 25. Memory Map Registers

Addr.

(Hex)

Chip Configuration Registers

0x00

0x01 Chip ID

0x02 Chip Grade

Channel Index and Transfer Registers

0x05

0xFF Transfer Open Open Open Open Open Open Open Transfer 0x00

ADC Functions Registers

0x08

0x09 Global Clock

0x0B Clock Divide

Register Name

SPI Port Configuration

(Global)

Channel Index

Power Modes

(Global)

Bit 7 (MSB)

0 LSB first Soft reset 1 1 Soft reset LSB first 0 0x18

Open Open Speed Grade ID[4:3]

Open Open Open Open Open Open

Open Open

Open Open Open Open Open Open Open

Open Open Open Open Open Clock divide ratio

Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1

8-bit Chip ID[7:0] (AD6653 = 0x0E)

(default)

00 = 150 MSPS 01 = 125 MSPS

External powerdown pin function (global)

0 = pdwn 1 = stndby

Open Open Open

Open Open Open Open

Data Channel B

(default)

Internal power-down mode (local)

00 = normal operation 01 = full power-down 10 = standby 11 = normal operation

000 = divide by 1 001 = divide by 2 010 = divide by 3 011 = divide by 4 100 = divide by 5 101 = divide by 6 110 = divide by 7 111 = divide by 8

Bit 0 (LSB)

Data Channel A

(default)

Duty cycle stabilize (default)

Default Value (Hex)

0x0E

0x03

0x00

0x01

0x00

Default Notes/ Comments

The nibbles are mirrored so that LSB-first or MSB-first mode registers correctly, regardless of shift mode

Default is unique chip ID, different for each device; this is a read-only register

Speed grade ID used to differentiate devices; this is a read-only register

Bits are set to determine which device on chip receives the next write command; applies to local registers

Synchronously transfers data from the master shift register to the slave

Determines various generic modes of chip operation

Clock divide values other than 000 automatically activate duty cycle stabilization

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Addr. (Hex)

0x0D Test Mode

0x10

0x14 Output Mode

0x16

0x17

0x18 VREF Select

Digital Feature Control Registers 0x100 Sync Control

0x101

0x102

0x103

0x104

Register Name

(Local)

Offset Adjust

(Local)

Clock Phase Control

(Global)

DCO Output Delay

(Global)

fS/8 Output Mix Control (Global)

FIR Filter and Output Mode Control (Global)

Digital Filter Control (Global)

Fast Detect Control (Local)

Bit 7 (MSB)

Open Open

Open Open Offset adjust in LSBs from +31 to −32 (twos complement format) 0x00

Drive strength

0 V to 3.3 V CMOS or ANSI LVDS

1 V to 1.8 V CMOS or reduced LVDS (global)

Invert DCO clock

Open Open Open

Reference voltage

Signal monitor sync enable

Open Open fS/8 start state Open Open

Open Open Open Open FIR gain

Open Open Open Open

Open Open Open Open Fast Detect Mode Select[2:0]

Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1

Reset PN long sequence

Output type

0 = CMOS 1 = LVDS (global)

Open Open Open Open Input clock divider phase adjust

selection

00 = 1.25 V p-p

01 = 1.5 V p-p

10 = 1.75 V p-p

11 = 2.0 V p-p (default)

Half-band next sync only

Interleaved CMOS (global)

Open Open Open Open Open Open 0xC0

Half-band sync enable

Reset PN short sequence

Output enable bar (local)

NCO32 next sync only

Open Output test mode

Open

(delay = 2500 ps × register value/31)

NCO32 sync enable

0 = gain of 2

1 = gain of 1

Half-band decimation phase

Output invert (local)

DCO clock delay

00000 = 0 ps 00001 = 81 ps

00010 = 161 ps …

11110 = 2419 ps

11111 = 2500 ps

Clock divider next sync only

fS/8 output mix disable

Spectral reversal

000 = off (default) 001 = midscale short 010 = positive FS 011 = negative FS 100 = alternating

101 = PN long sequence 110 = PN short sequence 111 = one/zero word toggle

00 = offset binary 01 = twos complement 01 = gray code 11 = offset binary (local)

000 = no delay 001 = 1 input clock cycle

010 = 2 input clock cycles 011 = 3 input clock cycles 100 = 4 input clock cycles 101 = 5 input clock cycles 110 = 6 input clock cycles 111 = 7 input clock cycles

Clock divider sync enable

fS/8 next sync only

Complex output enable

High-pass/ low-pass select

checkerboard

Default Bit 0 (LSB)

Master sync enable

fS/8 sync enable

FIR filter enable

Open 0x01

Fast detect enable

Value

(Hex)

0x00

Default Notes/ Comments

When enabled, the test data is placed on the output pins in place of ADC output data

Configures the outputs and the format of the data

Allows selection of clock delays into the input clock divider

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Addr. (Hex)

0x105

0x106

0x107

0x108

0x109

0x10A

0x10B

0x10C

0x10D

0x10E

0x10F

0x110

0x111

0x112

Register Name

Coarse Upper Threshold (Local)

Fine Upper Threshold Register 0 (Local)

Fine Upper Threshold Register 1 (Local)

Fine Lower Threshold Register 0 (Local)

Fine Lower Threshold Register 1 (Local)

Increase Gain Dwell Time Register 0 (Local)

Increase Gain Dwell Time Register 1 (Local)

Signal Monitor DC Correction Control (Global)

Signal Monitor DC Value Channel A Register 0 (Global)

Signal Monitor DC Value Channel A Register 1 (Global)

Signal Monitor DC Value Channel B Register 0 (Global)

Signal Monitor DC Value Channel B Register 1 (Global)

Signal Monitor SPORT Control

(Global)

Signal Monitor Control (Global)

Default Bit 7 (MSB)

Open Open Open Open Open Coarse Upper Threshold[2:0] 0x00

Open Open Open Fine Upper Threshold[12:8] 0x00

Open Open Open Fine Lower Threshold[12:8] 0x00

Open

Open Open DC Value Channel A[13:8] Read only

Open Open DC Value Channel B[13:8] Read only

Open

Complex power calculation mode enable

Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1

Fine Upper Threshold[7:0] 0x00

Fine Lower Threshold[7:0] 0x00

Increase Gain Dwell Time[7:0] 0x00

Increase Gain Dwell Time[15:8] 0x00

DC correction freeze

RMS magnitude output enable

Open Open Open

Peak detector output enable

DC Correction Bandwidth(k:[3:0])

DC Value Channel A[7:0] Read only

DC Value Channel B[7:0] Read only

Threshold crossing output enable

SPORT SMI SCLK divide

00 = Undefined 01 = divide by 2 10 = divide by 4 11 = divide by 8

Signal monitor rms/ms select

0 = rms

Signal monitor mode 00 = rms/ms magnitude 01 = peak detector 10 = threshold crossing

11 = threshold crossing

DC correction for signal path enable

SPORT SMI SCLK sleep

Bit 0 (LSB)

DC correction for signal monitor enable

Signal monitor SPORT output enable

Signal monitor enable

Value (Hex)

0x00

0x04

0x00

Default Notes/ Comments

In ADC clock cycles

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Addr. (Hex)

0x113

0x114

0x115

0x116

0x117

0x118

0x119

0x11A

0x11B

0x11D

0x11E NCO

0x11F NCO

0x120 NCO

0x121 NCO

0x122

0x123

Register Name

Signal Monitor Period Register 0 (Global)

Signal Monitor Period Register 1 (Global)

Signal Monitor Period Register 2 (Global)

Signal Monitor Value Channel A Register 0 (Global)

Signal Monitor Value Channel A Register 1 (Global)

Signal Monitor Value Channel A Register 2 (Global)

Signal Monitor Value Channel B Register 0 (Global)

Signal Monitor Value Channel B Register 1 (global)

Signal Monitor Value Channel B Register 2 (Global)

NCO Control

(Global)

Frequency 0

Frequency 1

Frequency 2

Frequency 3 NCO Phase

Offset 0 NCO Phase

Offset 1

Default Bit 7 (MSB)

Open Open Open Open Signal Monitor Result Channel A[19:16] Read only

Open Open Open Open Signal Monitor Result Channel B[19:16] Read only

Open Open Open Open Open

Bit 6 Bit 5 Bit 4 Bit 3 Bit 2 Bit 1

1 = ms

Signal Monitor Period[7:0] 0x80

Signal Monitor Period[15:8] 0x00

Signal Monitor Period[23:16] 0x00

Signal Monitor Result Channel A[7:0] Read only

Signal Monitor Result Channel A[15:8] Read only

Signal Monitor Result Channel B[7:0] Read only

Signal Monitor Result Channel B[15:8] Read only

NCO32 phase

dither enable

NCO Frequency Value[7:0] 0x00

NCO Frequency Value[15:8] 0x00

NCO Frequency Value[23:16] 0x00

NCO Frequency Value[31:24] 0x00

NCO Phase Value[7:0] 0x00

NCO Phase Value[15:8] 0x00

NCO32 amplitude dither enable

Bit 0 (LSB)

NCO32 enable

Value

(Hex)

0x00

Default Notes/ Comments

In ADC clock cycles

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MEMORY MAP REGISTER DESCRIPTION

For more information on functions controlled in Register 0x00 to Register 0xFF, see Application Note AN-877, Interfacing to High Speed ADCs via SPI, at www.analog.com.

SYNC Control (Register 0x100)

Bit 7—Signal Monitor Sync Enable

Bit 7 enables the sync pulse from the external sync input to the signal monitor block. The sync signal is passed when Bit 7 and Bit 0 are high. This is continuous sync mode.

Bit 6—Half-Band Next Sync Only

If the master sync enable bit (Register 0x100, Bit 0) and the halfband sync enable bit (Register 0x100, Bit 5) are high, Bit 6 allows the NCO32 to synchronize following the first sync pulse it receives and ignore the rest. If Bit 6 is set, Bit 5 of Register 0x100 resets after this sync occurs.

Bit 5—Half-Band Sync Enable

Bit 5 gates the sync pulse to the half-band filter. When Bit 5 is set high, the sync signal causes the half-band to resynchronize, starting at the half-band decimation phase selected in Register 0x103, Bit 3. This sync is active only when the master sync enable bit (Register 0x100, Bit 0) is high. This is continuous sync mode.

Bit 4—NCO32 Next Sync Only

If the master sync enable bit (Register 0x100, Bit 0) and the NCO32 sync enable bit (Register 0x100, Bit 3) are high, Bit 4 allows the NCO32 to sync following the first sync pulse it receives and ignores the rest. Bit 3 of Register 0x100 resets after a sync occurs if Bit 4 is set.

Bit 3—NCO32 Sync Enable

Bit 3 gates the sync pulse to the 32-bit NCO. When this bit is set high, the sync signal causes the NCO to resynchronize, starting at the NCO phase offset value. This sync is active only when the master sync enable bit (Register 0x100, Bit 0) is high. This is continuous sync mode.

Bit 2—Clock Divider Next Sync Only

If the master sync enable bit (Register 0x100, Bit 0) and the clock divider sync enable bit (Register 0x100, Bit 1) are high, Bit 2 allows the clock divider to synchronize following the first sync pulse it receives and ignore the rest. Bit 1 of Register 0x100 resets after it synchronizes.

Bit 1—Clock Divider Sync Enable

Bit 1 gates the sync pulse to the clock divider. The sync signal is passed when Bit 1 and Bit 0 are high. This is continuous sync mode.

Bit 0—Master Sync Enable

Bit 0 must be high to enable any of the sync functions.

fS/8 Output Mix Control (Register 0x101)

Bits[7:6]—Reserved

Bits[5:4]—f

Bit 5 and Bit 4 set the starting phase of the fS/8 output mix.

/8 Start State

S

Bits[3:2]—Reserved

Bit 1—f

If the master sync enable bit (Register 0x100, Bit 0) and the fS/8 sync enable bit (Register 0x101, Bit 0) are high, Bit 1 allows the

/8 output mix to synchronize following the first sync pulse it

f

S

receives and ignore the rest. Bit 0 of Register 0x100 resets after it synchronizes.

/8 Next Sync Only

S

Bit 0—fS/8 Sync Enable

Bit 0 gates the sync pulse to the fS/8 output mix. This sync is active only when the master sync enable bit (Register 0x100, Bit 0) is high. This is continuous sync mode.

FIR Filter and Output Mode Control (Register 0x102)

Bits[7:4]—Reserved

Bit 3—FIR Gain

When Bit 3 is set high, the FIR filter path, if enabled, has a gain of 1. When Bit 3 set low, the FIR filter path has a gain of 2.

Bit 2—fS/8 Output Mix Disable

Bit 2 disables the fS/8 output mix when enabled. Bit 2 should be set along with Bit 1 to enable complex output mode.

Bit 1—Complex Output Mode Enable

Setting Bit 1 high enables complex output mode.

Bit 0—FIR Filter Enable

When set high, Bit 0 enables the FIR filter. When Bit 0 is cleared, the FIR filter is bypassed and shut down for power savings.

Digital Filter Control (Register 0x103)

Bits[7:4]—Reserved

Bit 3—Half-Band Decimation Phase

When set high, Bit 3 uses the alternate phase of the decimating half-band filter.

Bit 2—Spectral Reversal

Bit 2 enables the spectral reversal feature of the half-band filter.

Bit 1—High-Pass/Low-Pass Select

Bit 1 enables the high-pass mode of the half-band filter when set high. Setting this bit low enables the low-pass mode (default).

Bit 0—Reserved

Bit 0 reads back as a 1.

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Fast Detect Control (Register 0x104)

Bits[7:4]—Reserved

Bits[3:1]—Fast Detect Mode Select

Bits[3:1] set the mode of the fast detect output bits according to Table 17.

Bit 0—Fast Detect Enable

Bit 0 is used to enable the fast detect output pins. When the FD outputs are disabled, the outputs go into a high impedance state. In LVDS mode when the outputs are interleaved, the outputs go high-Z only if both channels are turned off (power-down/ standby/output disabled). If only one channel is turned off (power-down/standby/output disabled), the fast detect outputs repeat the data of the active channel.

Coarse Upper Threshold (Register 0x105)

Bits[7:3]—Reserved

Bits[2:0]—Coarse Upper Threshold

These bits set the level required to assert the coarse upper threshold indication (see Table 21).

Fine Upper Threshold (Register 0x106 and Register 0x107)

Register 0x107, Bits[7:5]—Reserved

Register 0x107, Bits[4:0]—Fine Upper Threshold Bits[12:8]

Register 0x106, Bits[7:0]—Fine Upper Threshold Bits[7:0]

These registers provide a fine upper limit threshold. The 13-bit value is compared with the 13-bit magnitude from the ADC block and, if the ADC magnitude exceeds this threshold value, the F_UT indicator is set.

Fine Lower Threshold (Register 0x108 and Register 0x109)

Register 0x109, Bits[7:5]—Reserved

Register 0x109, Bits[4:0]—Fine Lower Threshold Bits[12:8]

Register 0x108, Bits[7:0]—Fine Lower Threshold Bits[7:0]

These registers provide a fine lower limit threshold. This 13-bit value is compared with the 13-bit magnitude from the ADC block and, if the ADC magnitude is less than this threshold value, the F_LT indicator is set.

Increase Gain Dwell Time (Register 0x10A and Register 0x10B)

Register 0x10B, Bits[7:0]—Increase Gain Dwell Time Bits[15:8]

Register 0x10A, Bits[7:0]—Increase Gain Dwell Time Bits[7:0]

These register values set the minimum time in ADC sample clock cycles (after clock divider) that a signal needs to stay below the fine lower threshold limit before the F_LT and IG are asserted high.

Signal Monitor DC Correction Control (Register 0x10C)

Bit 7—Reserved

Bit 6—DC Correction Freeze

When Bit 6 is set high, the dc correction is no longer updated to the signal monitor block, which holds the last dc value calculated.

Bits[5:2]—DC Correction Bandwidth

Bits[5:2] set the averaging time of the signal monitor dc correction function. This 4-bit word sets the bandwidth of the correction block, according to the following equation:

f

−−

14k

CLK

×=

2__

BWCorrDC

where:

k is the 4 bit value programmed in Bits[5:2] of Register 0x10C

(values between 0 and 13 are valid for k; programming 14 or 15 provides the same result as programming 13).

f

is the AD6653 ADC sample rate in hertz (Hz).

CLK

π×

2

Bit 1—DC Correction for Signal Path Enable

Setting this bit high causes the output of the dc measurement block to be summed with the data in the signal path to remove the dc offset from the signal path.

Bit 0—DC Correction for Signal Monitor Enable

This bit enables the dc correction function in the signal monitor block. The dc correction is an averaging function that can be used by the signal monitor to remove dc offset in the signal. Removing this dc from the measurement allows a more accurate power reading.

Signal Monitor DC Value Channel A (Register 0x10D and Register 0x10E)

Register 0x10E, Bits[7:6]—Reserved

Register 0x10E, Bits[5:0]—DC Value Channel A[13:8]

Register 0x10D, Bits[7:0]—DC Value Channel A[7:0]

These read-only registers hold the latest dc offset value computed by the signal monitor for Channel A.

Signal Monitor DC Value Channel B (Register 0x10F and Register 0x110)

Register 0x110, Bits[7:6]—Reserved

Register 0x110, Bits[5:0]—Channel B DC Value Bits[13:8]

Register 0x10F, Bits[7:0]—Channel B DC Value Bits [7:0]

These read-only registers hold the latest dc offset value computed by the signal monitor for Channel B.

Signal Monitor SPORT Control (Register 0x111)

Bit 7—Reserved

Bit 6—RMS/MS Magnitude Output Enable

Bit 6 enables the 20-bit rms or ms magnitude measurement as output on the SPORT.

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Bit 5—Peak Detector Output Enable

Bit 5 enables the 13-bit peak measurement as output on the SPORT.

Bit 4—Threshold Crossing Output Enable

Bit 4 enables the 13-bit threshold measurement as output on the SPORT.

Bits[3:2]—SPORT SMI SCLK Divide

The values of these bits set the SPORT SMI SCLK divide ratio from the input clock. A value of 0x01 sets divide-by-2 (default), a value of 0x10 sets divide-by-4, and a value of 0x11 sets divide-by-8.

Bit 1—SPORT SMI SCLK Sleep

Setting Bit 1 high causes the SMI SCLK to remain low when the signal monitor block has no data to transfer.

Bit 0—Signal Monitor SPORT Output Enable

When set, Bit 0 enables the signal monitor SPORT output to begin shifting out the result data from the signal monitor block.

Signal Monitor Control (Register 0x112)

Bit 7—Complex Power Calculation Mode Enable

This mode assumes I data is present on one channel and Q data is present on the alternate channel. The result reported is the complex power measured as

22

QI +

Bits[6:4]—Reserved

Bit 3—Signal Monitor RMS/MS Select

Setting Bit 3 low selects rms power measurement mode. Setting Bit 3 high selects ms power measurement mode.

Bits[2:1]—Signal Monitor Mode

Bit 2 and Bit 1 set the mode of the signal monitor for data output to registers at Address 0x116 through Address 0x11B. Setting these bits to 0x00 selects rms/ms magnitudde output, setting these bits to 0x01 selects peak detector output, and setting 0x10 or 0x11 selects threshold crossing output.

Bit 0—Signal Monitor Enable

Setting Bit 0 high enables the signal monitor block.

Signal Monitor Period (Register 0x113 to Register 0x115)

Register 0x115 Bits 7:0—Signal Monitor Period[23:16]

Register 0x114 Bits 7:0—Signal Monitor Period[15:8]

Register 0x113 Bits 7:0—Signal Monitor Period[7:0]

This 24-bit value sets the number of clock cycles over which the signal monitor performs its operation. The minimum value for this register is 128 cycles (programmed values less than 128 revert to 128).

Signal Monitor Result Channel A (Register 0x116 to Register 0x118)

Register 0x118, Bits[7:4]—Reserved

Register 0x118, Bits[3:0]—Signal Monitor Result

hannel A[19:16]

C

Register 0x117, Bits[7:0]—Signal Monitor Result C

hannel A[15:8]

Register 0x116, Bits[7:0]—Signal Monitor Result C

hannel A[7:0]

This 20-bit value contains the power value calculated by the signal monitor block for Channel A. The content is dependent on the settings in Register 0x112, Bits[2:1].

Signal Monitor Result Channel B (Register 0x119 to Register 0x11B)

Register 0x11B, Bits[7:4]—Reserved

Register 0x11B, Bits[3:0]—Signal Monitor Result

hannel B[19:16]

C

Register 0x11A, Bits[7:0]—Signal Monitor Result C

hannel B[15:8]

Register 0x119, Bits[7:0]—Signal Monitor Result C

hannel B[7:0]

This 20-bit value contains the power value calculated by the signal monitor block for Channel B. The content is dependent on the settings in Register 0x112, Bits[2:1].

NCO Control (Register 0x11D)

Bits[7:3]—Reserved

Bit 2—NCO32 Phase Dither Enable

When Bit 2 is set, phase dither in the NCO is enabled. When Bit 2 is cleared, phase dither is disabled.

Bit 1—NCO32 Amplitude Dither Enable

When Bit 1 is set, amplitude dither in the NCO is enabled. When Bit 1 is cleared, amplitude dither is disabled.

Bit 0—NCO32 Enable

When Bit 0 is set, this bit enables the 32-bit NCO operating at the frequency programmed into the NCO frequency register. When Bit 0 is cleared, the NCO is bypassed and shuts down for power savings.

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NCO Frequency (Register 0x11E to Register 0x121) NCO Phase Offset (Register 0x122 and Register 0x123)

Register 0x11E, Bits[7:0]—NCO Frequency Value[7:0] Register 0x122, Bits[7:0]—NCO Phase Value[7:0]

Register 0x11F, Bits [7:0]—NCO Frequency Value[15:8] Register 0x123, Bits[7:0]—NCO Phase Value[15:8]

Register 0x120, Bits[7:0]—NCO Frequency Value[23:16]

Register 0x121, Bits[7:0]—NCO Frequency Value[31:24]

This 32-bit value is used to program the NCO tuning frequency.

The frequency value to be programmed is given by the following

uation:

eq

ffMod

f

CLK

),(

NCO_FREQ

where:

O_FREQ is a 32-bit twos complement number representing

NC

the NCO frequency register. f is the desired carrier frequency in hertz (Hz).

is the AD6653 ADC clock rate in hertz (Hz).

f

CLK

32

2

×=

The 16-bit value programmed into the NCO phase value register is loaded into the NCO block each time the NCO is started or when an NCO SYNC signal is received. This process allows the NCO to be started with a known nonzero phase.

Use the following equation to calculate the NCO phase offset value:

NCO_PH

where:

O_PHASE is a decimal number equal to the 16-bit binary

NC

number to be programmed at Register 0x122 and Register 0x123.

PHASE is the desired NCO phase in degrees.

ASE = 2

16

× PHASE/360

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AD6653

–

R

–

R

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APPLICATIONS INFORMATION

DESIGN GUIDELINES

Before starting system-level design and layout of the AD6653, it is recommended that the designer become familiar with these guidelines, which discuss the special circuit connections and layout requirements needed for certain pins.

Power and Ground Recommendations

When connecting power to the AD6653, it is recommended that two separate 1.8 V supplies be used: one supply should be used for analog (AVDD) and digital (DVDD), and a separate supply should be used for the digital outputs (DRVDD). The AVDD and DVDD supplies, while derived from the same source, should be isolated with a ferrite bead or filter choke and separate decoupling capacitors. The designer can employ several different decoupling capacitors to cover both high and low frequencies. These capacitors should be located close to the point of entry at the PC board level and close to the pins of the part with minimal trace length.

A single PCB ground plane should be sufficient when using the AD6653. With proper decoupling and smart partitioning of the PCB analog, digital, and clock sections, optimum performance is easily achieved.

fS/2 Spurious

Because the AD6653 output data rate is at one-half the sampling frequency, there is significant f part. If this f that this f

/2 spur falls in band, care must be taken to ensure

S

/2 energy does not couple into either the clock circuit

S

or the analog inputs of the AD6653. When f in this fashion, it appears as a spurious tone reflected around f 3f

/4, 5fS/4, and so on. For example, in a 125 MSPS sampling

S

application with a 90 MHz single-tone analog input, this energy generates a tone at 97.5 MHz. In this example, the center of the Nyquist zone is 93.75 MHz; therefore, the 90 MHz input signal is

3.75 MHz from the center of the Nyquist zone. As a result, the f spurious tone appears at 97.5 MHz, or 3.75 MHz above the center of the Nyquist zone. These frequencies are then tuned by the NCOs before being output by the AD6653.

Depending on the relationship of the IF frequency to the center of the Nyquist zone, this spurious tone may or may not exist in the AD6653 output band. Some residual f the AD6653, and the level of this spur is typically below the level of the harmonics at clock rates of 125 MSPS and below. Figure 80 shows a plot of the f frequency for the AD6653-125. At sampling rates above 125 MSPS, the f

/2 spur level increases and is at a higher level

S

than the worst harmonic as shown in Figure 81, which shows the AD6653-150 f

/2 levels.

S

/2 energy in the outputs of the

S

/2 energy is coupled

S

/2 energy is present in

S

/2 spur level vs. analog input

S

/2

/4,

For the specifications provided in Table 2, the f band, is excluded from the SNR values. It is treated as a harmonic, in terms of SNR. The f

/2 level is included in the

S

SFDR and worst other specifications.

60

–70

–80

–90

/2 SPUR (dBFS )

S

AND f

–100

SFD

–110

–120

Figure 80. AD6653-125 SFDR and f

60

–70

–80

–90

/2 SPUR (dBFS )

S

AND f

–100

SFD

–110

–120

Figure 81. AD6653-150 SFDR and f

–SFDR

fS/2 SPUR

0 50 100 150 200 250 350300 400 450 500

with DRVDD = 1.8 V Parallel CMOS Output Mode

0 50 100 150 200 250 350300 400 450 500

with DRVDD = 1.8 V Parallel CMOS Output Mode

INPUT FREQUENCY (MHz)

–SFDR

/2 SPUR

f

S

INPUT FRE QUENCY (MHz)

/2 Spurious Level vs. Input Frequency (fIN)

S

/2 Spurious Level vs. Input Frequency (fIN)

S

Operating the part with a 1.8 V DRVDD voltage rather than 3.3 V DRVDD lowers the f

/2 spur. In addition, using LVDS, CMOS

S

interleaved, or CMOS IQ output modes also reduces the f spurious level.

LVDS Operation

The AD6653 defaults to CMOS output mode on power-up. If LVDS operation is desired, this mode must be programmed using the SPI configuration registers after power-up. When the AD6653 powers up in CMOS mode with LVDS termination resistors (100 Ω) on the outputs, the DRVDD current can be higher than the typical value until the part is placed in LVDS mode. This additional DRVDD current does not cause damage to the AD6653, but it should be taken into account when considering the maximum DRVDD current for the part.

/2 spur, if in

S

06708-083

06708-084

/2

S

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To avoid this additional DRVDD current, the AD6653 outputs can be disabled at power-up by taking the OEB pin high. After the part is placed into LVDS mode via the SPI port, the OEB pin can be taken low to enable the outputs.

Exposed Paddle Thermal Heat Slug Recommendations

It is mandatory that the exposed paddle on the underside of the ADC be connected to analog ground (AGND) to achieve the best electrical and thermal performance. A continuous, exposed (no solder mask), copper plane on the PCB should mate to the AD6653 exposed paddle, Pin 0.

The copper plane should have several vias to achieve the lowest p

ossible resistive thermal path for heat dissipation to flow through the bottom of the PCB. These vias should be filled or plugged with nonconductive epoxy.

To maximize the coverage and adhesion between the ADC and

he PCB, a silkscreen should be overlaid to partition the continuous

t plane on the PCB into several uniform sections. This provides several tie points between the ADC and the PCB during the reflow process. Using one continuous plane with no partitions guarantees only one tie point between the ADC and the PCB. See the evaluation board for a PCB layout example. For detailed information about packaging and PCB layout of chip scale packages, refer to Application Note AN-772, A Design and

Manufacturing Guide for the Lead Frame Chip Scale Package (LFCSP) (see

www.analog.com).

CML

The CML pin should be decoupled to ground with a 0.1 F capacitor, as shown in

RBIAS

The AD6653 requires that a 10 kΩ resistor be placed between the RBIAS pin and ground. This resistor sets the master current reference of the ADC core and should have at least a 1% tolerance.

Reference Decoupling

The VREF pin should be externally decoupled to ground with a low ESR, 1.0 F capacitor in parallel with a low ESR, 0.1 F ceramic capacitor.

SPI Port

The SPI port should not be active during periods when the full dynamic performance of the converter is required. Because the SCLK, CSB, and SDIO signals are typically asynchronous to the ADC clock, noise from these signals can degrade converter performance. If the on-board SPI bus is used for other devices, it may be necessary to provide buffers between this bus and the AD6653 to keep these signals from transitioning at the converter inputs during critical sampling periods.

Figure 48.

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EVALUATION BOARD

The AD6653 evaluation board provides all of the support circuitry required to operate the ADC in its various modes and configurations. The converter can be driven differentially through a double balun configuration (default) or optionally through the

AD8352 differential driver. The ADC can also be driven in a single-ended f

ashion. Separate power pins are provided to isolate the DUT

from the AD8352 drive circuitry. Each input configuration can

e selected by proper connection of various components (see

b Figure 83 to Figure 92). Figure 82 shows the typical bench

acterization setup used to evaluate the ac performance of

char the AD6653.

It is critical that the signal sources used for the analog input and

ck have very low phase noise (<<1 ps rms jitter) to realize the

clo optimum performance of the converter. Proper filtering of the analog input signal to remove harmonics and lower the integrated or broadband noise at the input is also necessary to achieve the specified noise performance.

See Figure 83 to Figure 100 for the complete schematics and

yout diagrams that demonstrate the routing and grounding

la techniques that should be applied at the system level.

POWER SUPPLIES

This evaluation board comes with a wall-mountable switching power supply that provides a 6 V, 2 A maximum output. Connect the supply to the rated 100 V ac to 240 V ac wall outlet at 47 Hz to 63 Hz. The output of the supply is a 2.1 mm inner diameter circular jack that connects to the PCB at J16. Once on the PC board, the 6 V supply is fused and conditioned before connection to six low dropout linear regulators that supply the proper bias to each of the various sections on the board.

WALL OUTLET 100V TO 240V AC 47Hz TO 63Hz

6V DC

2A MAX

SWITCHING

POWER

SUPPLY

ROHDE & SCHWARZ,

SMA100A, 2V p-p SIG NAL SYNTHESIZER

ROHDE & SCHWARZ,

SMA100A, 2V p-p SIG NAL SYNTHESIZER

ROHDE & SCHWARZ,

SMA100A, 2V p-p SI GNAL SYNTHESIZER

BAND-PASS

FILTER

BAND-PASS

FILTER

AINA

AINB

CLK

5.0V

–+

GND

1.8V

GND

AVDD IN

AMP VDD

AD6653

EVALUATIO N BOARD

Figure 82. Evaluation Board Connection

–+–+

3.3V

GND

External supplies can be used to operate the evaluation board

y removing L1, L3, L4, and L13 to disconnect the voltage

b regulators supplied from the switching power supply. This enables the user to individually bias each section of the board. Use P3 and P4 to connect a different supply for each section. At least one 1.8 V supply is needed with a 1 A current capability for AVDD and DVDD; a separate 1.8 V to 3.3 V supply is recommended for DRVDD. To operate the evaluation board using the AD8352 option, a separate 5.0 V supply (AMP VDD) with a

1 A current capability is needed. To operate the evaluation board using the alternate SPI options, a separate 3.3 V analog supply (VS) is needed, in addition to the other supplies. The 3.3 V supply (VS) should have a 1 A current capability, as well. Solder Jumper SJ35 allows the user to separate AVDD and DVDD, if desired.

INPUT SIGNALS

When connecting the clock and analog source, use clean signal generators with low phase noise, such as the Rohde & Schwarz SMA100A signal generators or the equivalent. Use 1 m long, shielded, RG-58, 50 Ω coaxial cable for making connections to the evaluation board. Enter the desired frequency and amplitude for the ADC. The AD6653 evaluation board from Analog Devices, Inc., can accept a ~2.8 V p-p or 13 dBm sine wave input for the clock. When connecting the analog input source, it is recommended that a multipole, narrow-band, band-pass filter with 50 Ω terminations be used. Band-pass filters of this type are available from TTE, Allen Avionics, and K&L Microwave, Inc. Connect the filter directly to the evaluation board, if possible.

OUTPUT SIGNALS

The parallel CMOS outputs interface directly with the Analog Devices standard ADC data capture board (HSC-ADC-EVALCZ). For more information on the ADC data capture boards and their optional settings, see www.analog.com/FIFO.

–+

DRVDD IN

3.3V

GND

–+

VS

PARALLEL

3.3V

VCP

GND

12-BIT

CMOS

12-BIT

CMOS

SPI SPI

HSC-ADC-EVALCZ

FPGA BASED

DATA

CAPTURE BOARD

USB

CONNECTION

PC RUNNING

VISUAL ANALO G

AND SPI

CONTROLL ER

SOFTWARE

6708-108

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DEFAULT OPERATION AND JUMPER SELECTION SETTINGS

The following is a list of the default and optional settings or modes allowed on the AD6653 evaluation board.

POWER

Connect the switching power supply that is provided in the evaluation kit between a rated 100 V ac to 240 V ac wall outlet at 47 Hz to 63 Hz and P500.

VIN

The evaluation board is set up for a double balun configuration analog input with optimum 50 Ω impedance matching from 70 MHz to 200 MHz. For more bandwidth response, the differential capacitor across the analog inputs can be changed or removed (see Ta b le 1 3 ). The common mode of the analog inputs

developed from the center tap of the transformer via the CML

is pin of the ADC (see the

VREF

VREF is set to 1.0 V by tying the SENSE pin to ground by adding a jumper on Header J5 (Pin 1 to Pin 2). This causes the ADC to operate in 2.0 V p-p full-scale range. To place the ADC in 1.0 V p-p mode (VREF = 0.5 V), a jumper should be placed on Header J4. A separate external reference option is also included on the evaluation board. To use an external reference, connect J6 (Pin 1 to Pin 2) and provide an external reference at TP5. Proper use of the VREF options is detailed in the

RBIAS

RBIAS requires a 10 kΩ resistor (R503) to ground and is used to

et the ADC core bias current.

s

CLOCK

The default clock input circuitry is derived from a simple baluncoupled circuit using a high bandwidth 1:1 impedance ratio balun (T5) that adds a very low amount of jitter to the clock path. The clock input is 50 Ω terminated and ac-coupled to handle singleended sine wave inputs. The transformer converts the single-ended input to a differential signal that is clipped before entering the ADC clock inputs. When the AD6653 input clock divider is utilized, clock frequencies up to 625 MHz can be input into the evaluation board through Connector S5.

PDWN

To enable the power-down feature, connect J7, shorting the PDWN pin to AVDD.

Analog Input Considerations section).

Volt age R ef e ren c e section.

CSB

The CSB pin is internally pulled up, setting the chip into external pin mode, to ignore the SDIO and SCLK information. To connect the control of the CSB pin to the SPI circuitry on the evaluation board, connect J21, Pin 1 to J21, Pin 2.

SCLK/DFS

If the SPI port is in external pin mode, the SCLK/DFS pin sets the data format of the outputs. If the pin is left floating, the pin is internally pulled down, setting the default data format condition to offset binary. Connecting J2, Pin 1 to J2, Pin 2 sets the format to twos complement. If the SPI port is in serial pin mode, connecting J2, Pin 2 to J2, Pin 3 connects the SCLK pin to the on-board SPI circuitry (see the

Serial Port Interface (SPI) section).

SDIO/DCS

If the SPI port is in external pin mode, the SDIO/DCS pin sets the duty cycle stabilizer. If the pin is left floating, the pin is internally pulled up, setting the default condition to DCS enabled. To disable the DCS, connect J1, Pin 1 to J1, Pin 2. If the SPI port is in serial pin mode, connecting J1, Pin 2 to J1, Pin 3 connects the SDIO pin to the on-board SPI circuitry (see the Serial Port

nterface (SPI)

I

section).

ALTERNATIVE CLOCK CONFIGURATIONS

Two alternate clocking options are provided on the AD6653 evaluation board. The first option is to use an on-board crystal oscillator (Y1) to provide the clock input to the part. To enable this crystal, Resistor R8 (0 Ω) and Resistor R85 (10 kΩ) should be installed, and Resistor R82 and Resistor R30 should be removed.

A second clock option is to use a differential LVPECL clock to dr

ive the ADC input using the AD9516 (U2). When using this ive option, the AD9516 charge pump filter components need

dr to be populated (see Figure 87). Consult the AD9516 data sheet

r more information.

fo

To configure the clock input from S5 to drive the AD9516

ference input instead of directly driving the ADC, the

re

ollowing components need to be added, removed, and/or

f changed.

Remove R32, R33, R99, and R101 in the default

1. clock path.

Populate C78 and C79 with 0.001 µF capacitors and

2. R78 and R79 with 0 Ω resistors in the clock path.

In addition, unused AD9516 outputs (one LVDS and one LVPECL)

routed to optional Connector S8 through Connector S11 on

are the evaluation board.

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Remove C1, C17, C18, and C117 in the default analog

ALTERNATIVE ANALOG INPUT DRIVE CONFIGURATION

This section provides a brief description of the alternative analog input drive configuration using the AD8352. When usin

g this particular drive option, some additional components

need to be populated. For more details on the AD8352 differential

iver, including how it works and its optional pin settings,

dr consult the

To configure the analog input to drive the AD8352 instead of th

e default transformer option, the following components need to be added, removed, and/or changed for Channel A. For Channel B the corresponding components should be changed.

AD8352 data sheet.

1. input path.

Populate C8 and C9 with 0.1 µF capacitors in the analog

2. input path. To drive the AD8352 in the differential input mode, populate the T10 transformer; the R1, R37, R39, R126, and R127 resistors; and the C10, C11, and C125 capacitors.

3.

Populate the optional amplifier output path with the

desired components including an optional low-pass filter. Install 0 Ω resistors, R44 and R48. R43 and R47 should be increased (typically to 100 Ω) to increase to 200 Ω the output impedance seen by the

AD8352.

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SCHEMATICS

DNPDNP

C139

12PF

AMP-A

AMP+A

AVD

AVDD

06708-090

DNPDNP

L16

180NH

12

IND0603

C4

18PF

120NH

12

L14

IND0603

C12

C2

0.1U

10K OH M

R41

AMPVDD

BA

W1

R40

10K OH M

R37

0OHM

AMPVDD

C10

0.1U

0.001U

12

11

GND

VON 10

VOP

VCC

13

VCM

14

15

16

R38

Z1

ENB

VIP

DN P

RGN

RGP

RDP

3

2

1

100 OH M

R127

C125

.3PF

4.12K

R126

DN P

R36

12

L17

IND0603

120NH

12

L15

IND0603

C16

9

GND

VCC

AD8352

GND

VIN

RDN

4

DNP

180NH

DNP

0.001U

8

67

5

R39

C11

R49

0OHM

1

TP14

AMPVDD

C27

10U

C23

0.1U

C22

0.1U

0OHM

0.1U

AMP-A

R44

0OHM

C17

R50

0OHM

VIN- A

1

TP15

C5

R26

33 OH M

R43

R42

CML

0.1U

VIN+A

4.7PF

R27

33 OH M

57.6 OH M

R5

C3

0.1U

33 OH M

R47

0OHM

AMP+A

R48

0OHM

C18

0.1U

R35

F

T10

R54

PS

4

24.9 OH M

CML

0OHM

T7

R110

0OHM

5

4

PS

T2

F

654

ADT1_1W T

123

F

T1

ETC1-1-1 3

123

PS

4

123

ETC1-1-1 3

5

0OHM

R4

24.9 OH M

R29

123

R31

0OHM

ETC1-1-1 3

5

DEFAULT AMP L IFIER INPUT PATH

C8

0.1U

OPTIONALAMPLIFIERINPUTPATH

INA-

C9

0.1U

INA-

INA+

S1

0.1U R2

C47

0.1U

C117

R120

0OHM

1

2

AIN-

0OHM

C1

0.1U

R121

0OHM

57.6 OH M

R28

RES0402

1

S2

AIN+

INA+

57.6 OH M

R1

2

Figure 83. Evaluation Board Schematic, Channel A Analog Inputs

Rev. 0 | Page 58 of 80

AD6653

0

www.BDTIC.com/ADI

AVDD

R80

C29

12PF

C19

18PF

10

VON

Z2

RGN

AMP-B

DNPDNP

12

L21

IND0603IND0603

12

L19

C140

9

GND

VCC

AD8352

GND

VIN

RDN

413

180NH

120NH

DNP DNP

0.001U

8

67

5

AMPVDD

AMP+B

DNP

180NH

12

L20

DNP

120NH

12

L18

IND0603 IND0603

C46

0.001U

C24

0.1U

12

10KOHM

R53

AMPVDD

BA

R131

10KOHM

W2

11

GND

VOP

VCC

13

VCM

15

ENB

16 14

VIP

RGP

RDP

2

0OHM

1

TP16

R73

C62

10U

C61

0.1U

C60

0.1U

AMP-B

R94

0OHM

AVDD

VIN-B

R81

0OHM

1

TP17

C84

33OHM

R70

R96

VIN+B

4.7PF

R74

33OHM

57.6OHM

R72

C83

0.1U

33OHM

R71

0OHM

CML

AMP+B

R95

0OHM

6708-091

C82

321

F

T4

4

321

F

PS

4

5

C6

R123

1

S3

0.1U

0OHM

R69

T3

0.1U

INB+

0OHM

RES0402

57.6OHM

R52

2

IN+

100OHM

AMPVDD

R132

C38

R66

DNP

R133

0OHM

0.1U

0OHM

R129

C128

.3PF

4.12K

R128

DNP

R68

24.9OHM

R134

3

2

1

F

PS

4

5

T11

ETC1-1-13

R6

0OHM

C39

0.1U

24.9OHM

R135

R55

0OHM

DEFAULT AMPLIFIER INPUT PATH

OPTIONAL AMPLIFIER INPUT PATH

C30

0.1U

INB+

C31

0.1U

INB-

C7

0.1U

ETC1-1-13

PS

CML

R111

0OHM

ADT1_1WT

INB -

C28

R122

1

S4

5

456

T8

321

ETC1-1-13

0.1U

C51

R67

0.1U

0OHM

RES0402

0OHM

57.6OHM

R51

2

IN-

Figure 84. Evaluation Board Schematic, Channel B Analog Inputs

Rev. 0 | Page 59 of 80

AD6653

www.BDTIC.com/ADI

CLK+

24.9OHM

C20

0.1U

R78

0OHM

ALTCLK+

VS

0.1U

C145

C64

R99

OPT_CLK+

C78

0.001U

456

C56

0.1U 321

OPT_CLK+

0.001U

R83

0OHM

R32

0OHM

R33

0OHM

T9

C63

1

ADT1_1WT

5

0.001U

3

2

F

T5

PS

ETC1-1-13

4

TP2

21

R84

DNP

R34

R101

OPT_CLK-

C79

0.001U

OPT_CLK-

C77

C94

0.001U

24.9OHM

R79

0OHM

ALTCLK-

06708-092

CLK -

C21

0.1U

10KOHM

R85

10KOHM

R82

R8

R90

0OHM

R30

1

2

S5

SMA200UP

ENC

R3

0OHM

57.6OHM

S6

SMA200UP

57.6OHM

R7

1

2

ENC\

Figure 85. Evaluation Board Schematic, DUT Clock Input

Rev. 0 | Page 60 of 80

AD6653

www.BDTIC.com/ADI

LVDS

OUTPU T

S8

1

S9

2

1

100 OH M

S10

1

LVPEC L

OUTPU T

S11

2

1

06708-093

R75

C88

0.1U

C87

0.1U

C85

0.1U

C86

0.1U

SYNC

200

C141

0.001U

100 OH M

R9

OUT6P

OUT6N

VS

49

5031

51

52

OUT1B

53

OUT1

54

55

VS_OUT_DR

4.12K

VS

R12

AGND

VS

5.1K

R11

OUT0B

56

OUT0

57

VS_REF

58

59

60

61

62

63

REFINB

64

REFIN

TP8

1

48

OUT6

TO ADC

LVPEC L

ALTCLK-

ALTCLK+

200

R86

200

R88

AGNDCP

44

43

46

47

OUT7

OUT6B

42

45

OUT2

OUT2B

OUT7B

GND_ESD

VS

VS_OUT_DR

41

VS_OUT23_DRV

AGND

37

40

39

OUT3

OUT3B

35

38

36

OUT9

OUT9B

VS_OUT23_DIV

GND_OUT89_DIV

VS_OUT67_1 VS_OUT67_2

VS_OUT01_DIV

VS_OUT01_DRV

U2

RSET_CLOCK

GND_REF

AD9516_64LFCSP

VS_PRESCALER VS_PLL_2 CP_RSET

R91

200

R92R125

33

34

OUT8

OUT8B

VS

PAD

VS_OUT89_2

32

VS_OUT89_1 VS_OUT45_DIV

30

OUT5B

29

OUT5

28

VS_OUT45_DRV

27

OUT4B

26

OUT4

25

PDB

24

RESETB

23

SDIO

22

SDO

21

NC4

20

NC3

19

NC2

18

CSB

17

VS_OUT_DR

PDB

RESETB

SDI

SDO

CSB_2

OPT_CL K +

OPT_CL K -

VS_PLL_1

1

VS

CP

LD

REFMO N

2

STATU S

VCP

3

6

4

5

STATU S

REFMON

VCP

1

TEST

TP19

R10

TEST

TP18

0OHM

C104

0.1U

LD

1

TEST

TP20

BYPASS_LDO

LF

REF_SEL

SYNCB

9

8

7

LF

SYNCB

REF_SEL

CLK

NC1

VS_CLK_DIST

VS_VCO

12

10

11

BYPASS_LDO

VS

C80

18PF

SCLK

CLKB

15

16

13

14

SCLK

0.1U

C143

C142

49.9 OH M

R89

R124

0OHM

RES0402

0OHM

VCXO_CLK+

1

2

S7

AD9516

CLK IN

Figure 86. Evaluation Board Schematic, Optional AD9516 Clock Circuit

Rev. 0 | Page 61 of 80

C97

0.1U

C96

0.1U

VCP

C99

0.1U

C98

0.1U

C101

0.1U

C100

0.1U

RES0402

VCXO_CLK-

VS_OUT_D R

AD6653

www.BDTIC.com/ADI

RESET BSYNCBPD BREF_SE L

RES0402

10KOHM

VSVSVSVS

R105

RES0402

10KOHM

R103

RES0402

10KOHM

R102

RES0402

10KOHM

R100

VCP

RES0402

10KOHM

R107

RES0402

10KOHM

R106

VCXO_CLK-

VCXO_CLK+

R139

R114

0OHM

RES0402

10KOHM

R109

RES0402

10KOHM

R108

06708-094

SYNC

24.9OHM

R87

R104

0OHM

RES0402

1

R46

33OHM

RES0402

4

5

Y1

Y2

VCC

NL27WZ04

A1

GND

A2

3

2

LD

C25

0.1U

RES0603

57.6OHM

R45

1

2

S12

SMA200UP

R116

AC

TP1

C26

0.1U

6

U3

1

VS

200

R76

VS

6

VCC

OSCVECTRON_VS500

FREQ_CTRL_V

1

LF

0OHM

RES0402

Charge Pump Filter

4

5

OUT2

OUT1

VS-500

U25

GND

OUT_DISABLE

3

2

0OHM

RES0402

C92

SEL

VALVALVAL

C144

SEL

C91

SEL

C89

R93

VAL

C90

SEL

VAL

R98

SYNC

R136 R137 R97 R117

Figure 87. Evaluation Board Schematic, Optional AD9516 Loop Filter/VCO and SYNC Input

Rev. 0 | Page 62 of 80

BYPASS_LDO

AD6653

3

www.BDTIC.com/ADI

D11B

D10B

D9B

D8B

D7B

D5B

D3A

1

D4A

D5A

D6A

D7A

D8A

D9A

D10A

D11A

8765432

RPAK8

8765432

22ohm

9

10111213141516

0.1U

C36

C35

0.001U

DRVD D

TP3

1

RES0402

R115

0OHM

RPAK8

22ohm

FD0A FD1A FD2A FD3A PWR_SD

PWR_SCLK PWR_SDF

R62

16

1

15

2

14

3

13

4

12

5

11

6

10

7

9

8

RES0402

R113

0OHM

R112

DVDD

22ohm

1

RPAK8

R61

17 18

20 21

24 25

29 30 31 32

D2A

D3A D4A D5A DRGND1 DRVDD1 D6A D7A DVDD1 D8A D9A D10A D11A_MSB_ FD0A FD1A FD2A FD3A

SMI_SDO/OEB

9

10111213141516

D1A

D0A_LSB

SMI_SCLK/PDW N

SMI_SDF S

343533

13591460156116

NC

AVDD1

36

1

NC

DCOA

VIN+A

VIN-A

3744384339

1

8765432

RPAK8

R60

22ohm

9

111012

D9B

D10B

DCOB

D11B_MSB

AD6653

CML

RBIAS

SENSE

VREF

41

40

DCOB

DCOA

SPARE

SPARE4D0A

D1A

D2A

D6B

10111213141516

6267278289

D8B

D7B

VIN+B

VIN-B

42

D4B

1

D0B

D1B

D2B

D3B

1

R59

0.001U

C33

2193224235

1

D4B

D5B

D6B

DRVDD

64

DRGND

63

D3B

62

D2B D1B

D0B_LSB

NC

58

NC

57

DVDD2

56

FD3B

55

FD2B

54

FD1B

53

FD0B

52

SYNC

51

SPI_CSB

50

CLK-

49

CLK+

U1

AVDD2

AVDD3

SPI_SCLK/DFS

SPI_SDIO/DCS

45

46

48

47

432

1

RPAK4

22ohm

567

8

0.1U

C34

R58

DRVD D

DVDD

SYNC

SPI_CSB CLKCLK+

FD1B

FD2B

FD3B

SPARE1

SPARE2

8765432

9

10111213141516

FD0B

06708-095

1

RPAK8

R57

22ohm

CM L

VIN-A

AVDD

VIN+A

DRVDD

J7- INSTALL FOR PDWN

J8- INSTALLFOROUTPUTDISABLE

C14

0.1U

J5- INSTALLFORIVVREF/2VINPUTSPAN

J4- INSTALLFO R0.5V VREF /I VINPUTSPA N

J6- INSTALLFOREXTERNALREFERENCEMODE

C15

1U

Figure 88. Evaluation Board Schematic, DUT

VIN-B

VIN+B

AVDD

SPI_SDIO

SPI_SCLK

C137

C120

0.1U

C32

RES0402

R63

R64

0OHM

10KOHM

1

TP6

AVDD

TP5

1

C40

0.1U

C126

0.001U

C127

0.001U

DVD D

0.001U

C121

0.1U

C109

0.1U

C122

0.001U

AVD D

Rev. 0 | Page 63 of 80

AD6653

www.BDTIC.com/ADI

C71

0.1U

C76

0.1U

C70

0.1U

C75

0.1U

C69

0.1U

C68

0.1U

C67

0.1U

C66

0.1U

C65

0.1U

V_DIG

J12

A1

A2

B2

C1

C2

D1

D2

TYCO_HM-ZD

RES0402

10KOHM

VS

R118

R119

0OHM

1

RES0402

TEST

TP21

RESETB

C74

0.1U

C73

0.1U

C72

0.1U

V_DIG

DG10

DG9

DG8

DG7

DG6

DG5

DG4

DG3

DG2

DG1

BG10

BG9

BG8

BG7

BG6

BG5

BG4

BG3

BG2

BG1

A3

A4

A5

A6

A7

A8

A9

B3

B4

B5

B6

B7

C3

C4

D3

R143

0OHM

SDO

VS

C5

D4

D5

A10

B10

R142

0OHM

SDFS_OUT

RES0402

SDI

R141

0OHM

SCLK_OUT

RES0402

10KOHM

R140

R144

0OHM

B8

C6

C7

D6

VAL R130R77

R145

RES0402

C8

D7

D8

1

TEST

TP24

TP23

SYNC

0OHM

RES0402

B1

B9

C9

D9

C10

D10

SDO_OUT

1

TEST

TP22

100OHM

OUT6P

06708-096

OUT6N

CSB_2

DG10

DG9

DG8

DG7

DG6

DG5

DG4

DG3

DG2

DG1

BG10

BG9

BG8

BG7

BG6

BG5

BG4

BG3

BG2

BG1

A2

A3

A4

A5

A6

A7

A8

A9

B1

C9

CHANNELA

C10

D10

SCLK_OU T

SDFS_OUT

DIGITAL/HSC-ADC-EVALCZ INTERFACE

24

U15

25

PWR_SCL K

PWR_SDF S

B8

B9

C7

C8

D8

D9

V_DI G

SDO_OUT

V_DIG

FD3A

PWR_SDO

B6

B7

D7

FD2A

FD1A

FD0A

C5

C6

D6

A10

B10

D8A

D9A

D10A

D11A

V_DIG

B4

B5

C3

C4

D4

D5

1234567891011121314151617181920212223

U16

74VCX162244MTD

4847464544434241403938373635343332313029282726

D6A

D7A

B2

B3

C1

C2

D1

D2

D3

V_DIG

24

25

D1A

D2A

D3A

D4A

D5A

V_DIG

Figure 89. Evaluation Board Schema

CSB

SCLK

J10

A1

B1

C10

D10

TYCO_HM-ZD

OUT6N

D0A

D11B

DCOB

DCOA

SPARE3

SPARE4

A8

A9

B9

C8

C9

D8

D9

1234567891011121314151617181920212223

4847464544434241403938373635343332313029282726

V_DIG

D6B

D7B

D8B

D9B

D10B

BG6

BG5

BG4

BG3

BG2

BG1

A7

B7

B8

C6

C7

D6

D7

24

U17

74VCX162244MTD

25

DG3

DG2

DG1

BG10

BG9

BG8

BG7

A5

A6

B6

C5

D5

A10

B10

V_DI G

D1B

D2B

D3B

D4B

D5B

V_DIG

DG10

DG9

DG8

DG7

DG6

DG5

DG4

J11

CHANNELB

A1

A2

A3

A4

B4

B5

C3

C4

D4

D0B

FD2B

FD3B

SPARE1

SPARE2

B2

B3

C1

C2

D1

D2

D3

V_DI G

FD0B

FD1B

V_DIG

TYCO_HM-ZD

OUT6P

1234567891011121314151617181920212223

74VCX162244MTD

4847464544434241403938373635343332313029282726

tic, Digital Output Interface

Rev. 0 | Page 64 of 80

AD6653

T

www.BDTIC.com/ADI

JUMPERPINS1 TO 2 FORDCSENABLE

JUMPERPINS1 TO 2 FORTWOSCOMPLIMENTOUTPU

J1- JUMPERPINS2 TO3 FORSPIOPERATION

J2- JUMPERPINS2 TO3 FORSPIOPERATION

J21- INSTALLJUMPERFORSPIOPERATION

V_DI G

SPI_SCLK

SPI_SDIO

1

J1

3

1

J2

3

R22

R23

100KOHM

RES0603

SPI_CSBV_DIG

100KOHM

RES0603

06708-097

V_DIG

R20

R19

U7

SDI

VS

R21

1KOHM

RES0603

6

Y1

RES0603

A1

1

R18

RES0603

V_DIG

SDO

4

5

Y2

VCC

NC7WZ07P6X

GND

A2

3

2

C13

0.1U

10KOHM

RES0402

V_DI G

C81

0.1U

U8

R24

10KOHM

RES0402

CSB_2

CSB

SDI

SDO

SCLK

Figure 90. Evaluation Board Schematic, SPI Circuitry

R17

100KOHM

RES0603

V_DI G

456

Y1

Y2

VCC

NC7WZ16P6X

A1

GND

A2

321

SCLK

CSB

RES0402

10KOHM

R65

Rev. 0 | Page 65 of 80

AD6653

N

www.BDTIC.com/ADI

DRVDDI

21

L4

10uh

IND1210

C45

AVDDI N

21

L3

10uh

IND1210

C43

1U

ADP3339

OUT

GND

4

1

PAD

VR3

IN

3

C42

1U

21

CR12

S2A_RECT

21

CR11

S2A_RECT

VR1

ADP3334

1U

C93

0.001U

76.8KOHM

R13R14

147KOHM

DRVDDSETTING

1

2

3

FB

GND

OUT

IN2

IN

8

7

5

OUT2

SD

6

C44

1U

147K

78.7K

94.0K

140K

107K

76.8K

3.3

1.8

2.5

DRVDD R13 R14

TP4

TP9

1

TP131TP121TP10

GNDTEST POINTS

06708-098

1

21

CR10

S2A_RECT

RES0603

AC

PWR_IN

21

CR8

SHOT_RECT

245

6

CG

CB

CG

F1

BIAS

PSG

1

3

CR7

21

S2A_RECT

F2

SMDC110F

C41

10U

2

J16

POWER_JACK

6V,2A MAX

3

1

POWERINPUT

261OHM

R16

AVDD

C57

0.1U

C52

L6

IND1210

10UH

12

SJ35

10U

BNX-016

AVDDIN

1P12P23P34P45P56

P3

DVDD

C103

0.1U

C102

10U

21

L9

IND1210

10UH

P6

PTIONALPOWERSUPPLYINPUTS

1

TP25

21

IND1210

DRVDDIN

P4

V_DI G

21

DRVDD

L11

10uh

IND1210

C58

0.1U

C53

10U

L10

10uh

VS

P1P2P3

C59

0.1U

C54

10U

VCP

P4

Figure 91. Evaluation Board Schematic, Power Supply

Rev. 0 | Page 66 of 80

AD6653

www.BDTIC.com/ADI

C105

AMPVDD

21

L1

IND1210

10UH

C130

1U

OUT

GND

4

1

PAD

ADP3339

IN

VR4

3

C129

1U

PWR_IN

VS_OUT_DR

21

L12

IND1210

10uh

VS

21

L13

IND1210

10uh

0.1U

C116

0.1U

C107

0.1U

C113

0.1U

C114

0.1U

C115

0.1U

C111

0.1U

C108

0.1U

C112

0.1U

C110

0.1U

VS

VCP

21

L8

IND1210

10UH

C131

1U

06708-099

SJ36

C134

1U

OUT

GND

4

1

PAD

ADP3339

IN

VR5

3

C133

1U

PWR_IN

OUT

4

PAD

ADP3339

IN

VR6

3

PWR_IN

SJ37

C95

0.001U

140KOHM

C136

1U

GND

1

C135

1U

VR2

ADP3334

R25

1

2

3

FB

5

GND

OUT

OUT2

6

SD

IN2

IN

8

7

C132

1U

PWR_IN

78.7KOHM

R15

C118

10U

VS

C124

10U

VS_OUT_DR

C119

10U

VCP

Power Supply ByPass Capacitors

Figure 92. Evaluation Board Schematic, Power Supply (Continued)

Rev. 0 | Page 67 of 80

AD6653

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EVALUATION BOARD LAYOUTS

Figure 93. Evaluation Board Layout, Primary Side

Rev. 0 | Page 68 of 80

06708-100

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Figure 94. Evaluation Board Layout, Ground Plane

Rev. 0 | Page 69 of 80

6708-101

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Figure 95. Evaluation Board Layout, Power Plane

Rev. 0 | Page 70 of 80

6708-102

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Figure 96. Evaluation Board Layout, Power Plane

Rev. 0 | Page 71 of 80

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Figure 97. Evaluation Board Layout, Ground Plane

Rev. 0 | Page 72 of 80

06708-104

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Figure 98. Evaluation Board Layout, Secondary Side (Mirrored Image)

Rev. 0 | Page 73 of 80

6708-105

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Figure 99. Evaluation Board Layout, Silkscreen, Primary Side

Rev. 0 | Page 74 of 80

06708-106

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Figure 100. Evaluation Board Layout, Silkscreen, Secondary Side

Rev. 0 | Page 75 of 80

6708-107

AD6653

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BILL OF MATERIALS

Table 26. Evaluation Board Bill of Materials (BOM)

Reference

gnator

Item Qty

1 1 AD6653CE_REVB PCB PCB Analog Devices 2 55 C1 to C3, C6, C7,

3 1 C80 18 pF, COG, 50 V, 5% ceramic

4 2 C5, C84 4.7 pF, COG, 50 V, 5% ceramic

5 10 C33, C35, C63,

6 13 C15, C42 to C45,

7 10 C27, C41, C52 to

8 1 CR5 Schottky diode HSMS2822, SOT23 SOT23 Avago Technologies HSMS-2822-BLKG

9 2 CR6, CR9 LED RED, SMT, 0603, SS-type LED0603 Panasonic LNJ208R8ARA

Desi

C13, C14, C17, C18, C20 to C26, C32, C57 to C61, C65 to C76, C81 to C83, C96 to C101, C103, C105, C107, C108, C110 to C116, C145

C93 to C95, C122, C126, C127, C137

C129 to C136

C54, C62, C102, C118, C119, C124

Description Package Manufacturer Mfg. Part Number

0.1 μF, 16 V ceramic capacitor,

capacitor, SMT 0402

0.001 μF, X7R, 25 V, 10% cer

1 μF, X5R, 25 V, 10% ceramic capacitor, SMT 0805

10 μF, X5R, 10 V, 10% ceramic capacitor,

SMT 0402

amic capacitor, SMT 0402

SMT 1206

1, 2

C0402SM Murata GRM155R71C104KA88D

C0402SM Murata GJM1555C1H180JB01J

C0402SM Murata GJM1555C1H4R7CB01J

C0402SM Murata GRM155R71H102KA01D

C0805 Murata GR4M219R61A105KC01D

C1206 Murata GRM31CR61C106KC31L

10 4 CR7, CR10 to CR12 50 V, 2 A diode DO_214AA Micro Commercial Components S2A-TP

11 1 CR8 30 V, 3 A diode DO_214AB Micro Commercial Components SK33-TP

12 1 F1 EMI filter FLTHMURATABNX01 Murata BNX016-01

13 1 F2 6.0 V, 3.0 A, trip current

14 2 J1, J2 3-pin, male, single row,

15 9 J4 to J9, J18, J19,

J21

16 3 J10 to J12 Interface connector TYCO_HM_ZD Tyco 6469169-1

17 1 J14 8-pin, male, double row,

18 1 J16 DC power jack connector PWR_JACK1 Cui Stack PJ-002A

19 10 L1, L3, L4, L6, L8

to L13

20 1 P3 6-terminal connector PTMICRO6 Weiland Electric, Inc. Z5.531.3625.0

21 1 P4 4-terminal connector PTMICRO4 Weiland Electric, Inc. Z5.531.3425.0

22 3 R7, R30, R45 57.6 Ω, 0603, 1/10 W,

23 27 R2, R3 R4, R32,

R33, R42, R64, R67, R69, R90, R96, R99, R101, R104, R110 to R113, R115, R119, R121, R123, R141 to R145

24 1 R13 76.8 kΩ, 0603, 1/10 W, 1% resistor R0603 NIC Components NRC06F7682TRF

25 1 R25 140 kΩ, 0603, 1/10 W, 1% resistor R0603 NIC Components NRC06F1403TRF

26 1 R14 147 kΩ, 0603, 1/10 W, 1% resistor R0603 NIC Components NRC06F1473TRF

27 1 R15 78.7 kΩ, 0603, 1/10 W,

resettable fuse

straight header 2-pin, male, straight header HDR2 Samtec TWS-102-08-G-S

straight header

10 μH, 2 A bead core, 1210 1210 Panasonic EXC-CL3225U1

1% resistor 0 Ω, 1/16 W, 5% resistor R0402SM NIC Components NRC04ZOTRF

1% resistor

L1206 Tyco Raychem NANOSMDC150F-2

HDR3 Samtec TWS-1003-08-G-S

CNBERG2X4H350LD Samtec TSW-104-08-T-D

R0603 NIC Components NRC06F57R6TRF

R0603 NIC Components NRC06F7872TRF

Rev. 0 | Page 76 of 80

AD6653

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Reference

Item Qty

28 1 R16 261 Ω, 0603, 1/10 W, 1% resistor R0603 NIC Components NRC06F2610TRF

29 3 R17, R22, R23 100 kΩ, 0603, 1/10 W, 1% resistor R0603 NIC Components NRC06F1003TRF

30 7 R18, R24, R63, R65,

31 3 R19, R21 1 kΩ, 0603, 1/10 W, 1% resistor R0603 NIC Components NRC06F1001TRF

32 9 R26, R27, R43,

33 5 R57, R59 to R62 22 Ω, 16-pin, 8-resistor,

34 1 R58 22 Ω, 8-pin, 4-resistor,

35 1 R76 200 Ω, 0402, 1/16 W,

36 4 S2, S3, S5 ,S12 SMA, inline, male,

37 1 SJ35 0 Ω, 1/8 W, 1% resistor SLDR_PAD2MUYLAR NIC Components NRC10ZOTRF

38 5 T1 to T5 Balun TRAN6B M/A-COM MABA-007159-000000

39 1 U1 IC, AD6653 LFCSP64-9X9-9E Analog Devices AD6653BCPZ

40 1 U2 Clock distribution, PLL IC LFCSP64-9X9 Analog Devices AD9516-4BCPZ

41 1 U3 Dual inverter IC SC70_6 Fairchild Semiconductor NC7WZ04P6X_NL

42 1 U7 Dual buffer IC,

43 1 U8 UHS dual buffer IC SC70_6 Fairchild Semiconductor NC7WZ16P6X_NL

44 3 U15 to U17 16-bit CMOS buffer IC TSOP48_8_1MM Fairchild Semiconductor 74VCX16244MTDX_NL

45 2 VR1, VR2 Adjustable regulator LFCSP8-3X3 Analog Devices ADP3334ACPZ

46 1 VR3 1.8 V high accuracy regulator SOT223-HS Analog Devices ADP3339AKCZ-1.8

47 1 VR4 5.0 V high accuracy regulator SOT223-HS Analog Devices ADP3339AKCZ-5.0

48 2 VR5, VR6 3.3 V high accuracy regulator SOT223-HS Analog Devices ADP3339AKCZ-3.3

49 1 Y1 Oscillator clock, VFAC3 OSC-CTS-CB3 Valpey Fisher VFAC3-BHL

50 2 Z1, Z2 High speed IC, op amp LFCSP16-3X3-PAD Analog Devices AD8352ACPZ

1

This bill of materials is RoHS compliant.

2

The bill of materials lists only those items that are normally installed in the default condition. Items that are not installed are not included in the BOM.

Designator

R82, R118, R140

R46, R47, R70, R71, R73, R74

Description Package

10 kΩ, 0402, 1/16 W, 1% resistor R0402SM NIC Components NRC04F1002TRF

33 Ω, 0402, 1/16 W, 5% resistor R0402SM NIC Components NRC04J330TRF

R_742 CTS Corporation 742C163220JPTR

resistor array

RES_ARRY CTS Corporation 742C083220JPTR

resistor array

R0402SM NIC Components NCR04F2000TRF

1% resistor

SMA_EDGE Emerson Network

coaxial connector

SC70_6 Fairchild Semiconductor NC7WZ07P6X_NL

open-drain circuits

Manufacturer Mfg. Part Number

142-0701-201

Power

Rev. 0 | Page 77 of 80

AD6653

www.BDTIC.com/ADI

OUTLINE DIMENSIONS

49

48

0.60 MAX

EXPOSED PAD

(BOTTOM VIEW)

PIN 1

64

INDICATOR

1

7.25

7.10 SQ

6.95

PIN 1

INDICATOR

9.00

BSC SQ

TOP VIEW

8.75

BSC SQ

0.60

MAX

0.50

BSC

1.00

0.85

0.80

SEATING

PLANE

12° MAX

0.50

0.40

0.30

0.80 MAX

0.65 TYP

0.30

0.23

0.18

COMPLIANT TO JEDEC S TANDARDS MO-220-VMMD-4

0.05 MAX

0.02 NOM

0.20 REF

33

32

7.50 REF

16

17

0.25 MIN

051007-C

Figure 101. 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ]

9

mm × 9 mm Body, Very Thin Quad

(CP-64-3)

Dimensions shown in millimeters

ORDERING GUIDE

Model Temperature Range Package Description Package Option

AD6653BCPZ-150 AD6653BCPZ-125 AD6653-125EBZ AD6653-150EBZ

1

Z = RoHS Compliant Part.

1

−40°C to +85°C 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ] CP-64-3

1

−40°C to +85°C 64-Lead Lead Frame Chip Scale Package [LFCSP_VQ] CP-64-3

1

Evaluation Board with AD6653 and Software

1

Evaluation Board with AD6653 and Software

Rev. 0 | Page 78 of 80

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NOTES

Rev. 0 | Page 79 of 80

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NOTES

Rev. 0 | Page 80 of 80

Datasheet AD6653 Datasheet (ANALOG DEVICES)

Specifications and Main Features

Frequently Asked Questions

User Manual

FEATURES

FUNCTIONAL BLOCK DIAGRAM

APPLICATIONS

PRODUCT HIGHLIGHTS

TABLE OF CONTENTS

REVISION HISTORY

GENERAL DESCRIPTION

SPECIFICATIONS

ADC DC SPECIFICATIONS

ADC AC SPECIFICATIONS

DIGITAL SPECIFICATIONS

SWITCHING SPECIFICATIONS

TIMING SPECIFICATIONS

Timing Diagrams

ABSOLUTE MAXIMUM RATINGS

THERMAL CHARACTERISTICS

ESD CAUTION

PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS

EQUIVALENT CIRCUITS

TYPICAL PERFORMANCE CHARACTERISTICS

THEORY OF OPERATION

ADC ARCHITECTURE

ANALOG INPUT CONSIDERATIONS

Input Common Mode

Differential Input Configurations

Single-Ended Input Configuration

VOLTAGE REFERENCE

Internal Reference Connection

CLOCK INPUT CONSIDERATIONS

External Reference Operation

Clock Input Options

Input Clock Divider

Clock Duty Cycle

Jitter Considerations

POWER DISSIPATION AND STANDBY MODE

DIGITAL OUTPUTS

Digital Output Enable Function (OEB)

Interleaved CMOS Mode

Timing

Data Clock Output (DCO)

DIGITAL DOWNCONVERTER

DOWNCONVERTER MODES

NUMERICALLY CONTROLLED OSCILLATOR (NCO)

HALF-BAND DECIMATING FILTER AND FIR FILTER

NUMERICALLY CONTROLLED OSCILLATOR (NCO)

FREQUENCY TRANSLATION

NCO SYNCHRONIZATION

PHASE OFFSET

NCO AMPLITUDE AND PHASE DITHER

DECIMATING HALF-BAND FILTER AND FIR FILTER

HALF-BAND FILTER COEFFICIENTS

HALF-BAND FILTER FEATURES

FIXED-COEFFICIENT FIR FILTER

SYNCHRONIZATION

COMBINED FILTER PERFORMANCE

FINAL NCO

ADC OVERRANGE AND GAIN CONTROL

FAST DETECT OVERVIEW

ADC fast magnitude

ADC OVERRANGE (OR)

GAIN SWITCHING

Coarse Upper Threshold (C_UT)

Fine Upper Threshold (F_UT)

Fine Lower Threshold (F_LT)

Increment Gain (IG) and Decrement Gain (DG)

SIGNAL MONITOR

PEAK DETECTOR MODE

RMS/MS MAGNITUDE MODE

THRESHOLD CROSSING MODE

ADDITIONAL CONTROL BITS

Signal Monitor Enable Bit

Complex Power Calculation Mode Enable Bit

DC CORRECTION

DC Correction Bandwidth

DC Correction Readback

DC Correction Freeze