Datasheet AD636 Datasheet (ANALOG DEVICES)

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Low Level, True RMS-to-DC Converter

FEATURES

True rms-to-dc conversion 200 mV full scale Laser-trimmed to high accuracy
0.5% maximum error (AD636K)
1.0% maximum error (AD636J)
Wide response capability
Computes rms of ac and dc signals 1 MHz, −3 dB bandwidth: V rms > 100 mV
Signal crest factor of 6 for 0.5% error dB output with 50 dB range Low power: 800 μA quiescent current Single or dual supply operation Monolithic integrated circuit Low cost

GENERAL DESCRIPTION

The AD636 is a low power monolithic IC that performs true rms-to-dc conversion on low level signals. It offers performance that is comparable or superior to that of hybrid and modular converters costing much more. The AD636 is specified for a signal range of 0 mV to 200 mV rms. Crest factors up to 6 can be accommodated with less than 0.5% additional error, allowing accurate measurement of complex input waveforms.
AD636

FUNCTIONAL BLOCK DIAGRAM

dB
R
L
I
OUT
BUF OUT
10k
Figure 1.
SQUARER
DIVIDER
CURRENT
MIRROR
10k
+V
S
BUF IN
V
IN
C
AV
ABSOLUTE
VALUE
AD636
at 200 mV rms. Therefore, no external trims are required to achieve full-rated accuracy.
The AD636 is available in two accuracy grades. The total error o
f the J-version is typically less than ±0.5 mV ± 1.0% of reading, while the total error of the AD636K is less than ±0.2 mV to ±0.5% of reading. Both versions are temperature rated for operation between 0°C and 70°C and available in 14-lead SBDIP and 10-lead TO-100 metal can.
00787-001
The low power supply current requirement of the AD636,
ically 800 A, is ideal for battery-powered portable
typ instruments. It operates from a wide range of dual and single power supplies, from ±2.5 V to ±16.5 V or from +5 V to +24 V. The input and output terminals are fully protected; the input signal can exceed the power supply with no damage to the device (allowing the presence of input signals in the absence of supply voltage), and the output buffer amplifier is short-circuit protected.
The AD636 includes an auxiliary dB output derived from an in
ternal circuit point that represents the logarithm of the rms output. The 0 dB reference level is set by an externally supplied current and can be selected to correspond to any input level from 0 dBm (774.6 mV) to −20 dBm (77.46 mV). Frequency response ranges from 1.2 MHz at 0 dBm to greater than 10 kHz at −50 dBm.
The AD636 is easy to use. The device is factory-trimmed at the wa
fer level for input and output offset, positive and negative
waveform symmetry (dc reversal error), and full-scale accuracy
Rev. D
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Anal og Devices for its use, nor for any infringements of patents or ot her rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
The AD636 computes the true root-mean-square of a complex ac (o
r ac plus dc) input signal and gives an equivalent dc output level. The true rms value of a waveform is a more useful quantity than the average rectified value because it is a measure of the power in the signal. The rms value of an ac-coupled signal is also its standard deviation.
The 200 mV full-scale range of the AD636 is compatible with ma
ny popular display-oriented ADCs. The low power supply current requirement permits use in battery-powered hand-held instruments. An averaging capacitor is the only external component required to perform measurements to the fully specified accuracy is. Its value optimizes the trade-off between low frequency accuracy, ripple, and settling time.
An optional on-chip amplifier acts as a buffer for the input or
e output signals. Used in the input, it provides accurate
th performance from standard 10 M input attenuators. As an output buffer, it sources up to 5 mA.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 ©2006 Analog Devices, Inc. All rights reserved.
AD636
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TABLE OF CONTENTS

Features.............................................................................................. 1
AD636 Principle of Operation................................................9
Functional Block Diagram ..............................................................1
General Description......................................................................... 1
Revision History ...............................................................................2
Specifications..................................................................................... 3
Absolute Maximum Ratings............................................................ 5
ESD Caution.................................................................................. 5
Pin Configurations and Function Descriptions........................... 6
Applying the AD636......................................................................... 7
Standard Connection................................................................... 7
Optional Trims for High Accuracy............................................ 7
Single-Supply Connection........................................................... 7
Choosing the Averaging Time Constant................................... 8
RMS Measurements..................................................................... 9

REVISION HISTORY

The AD636 Buffer Amplifier.......................................................9
Frequency Response .................................................................. 10
AC Measurement Accuracy and Crest Factor (CF)............... 10
A Complete AC Digital Voltmeter........................................... 11
A Low Power, High Input, Impedance dB Meter....................... 11
Circuit Description................................................................ 11
Performance Data .................................................................. 11
Frequency Response ±3 dBm............................................... 11
Calibration .............................................................................. 11
Outline Dimensions....................................................................... 13
Ordering Guide .......................................................................... 13
11/06—Rev. C to Rev. D
Changes to General Description .................................................... 1
Changes to Table 1............................................................................ 3
Changes to Ordering Guide.......................................................... 13
1/06—Rev B to Rev. C
pdated Format..................................................................Universal
U
Changes to Figure 1 and General Description ............................. 1
Deleted Metalization Photograph.................................................. 3
Added Pin Configuration and Function Description Section.... 6
Updated Outline Dimensions....................................................... 14
Changes to Ordering Guide.......................................................... 14
8/99—Rev A to Rev. B
Rev. D | Page 2 of 16
AD636
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SPECIFICATIONS

@ 25°C, +VS = +3 V, and −VS = –5 V, unless otherwise noted.
Table 1.
Model
TRANSFER FUNCTION
CONVERSION ACCURACY
Total Error, Internal Trim
vs. Temperature, 0°C to +70°C ±0.1 ± 0.01 ±0.1 ± 0.005
vs. Supply Voltage ±0.1 ± 0.01 ±0.1 ± 0.01
DC Reversal Error at 200 mV ±0.2 ±0.1 % of reading Total Error, External Trim2 ±0.3 ± 0.3 ± 0.1 ± 0.2
ERROR VS. CREST FACTOR4
Crest Factor 1 to 2 Specified Accuracy Specified Accuracy Crest Factor = 3 −0.2 −0.2 % of reading
Crest Factor = 6 −0.5 −0.5 % of reading AVERAGING TIME CONSTANT 25 25 ms/F of CAV INPUT CHARACTERISTICS
Signal Range, All Supplies
Continuous RMS Level 0 to 200 0 to 200 mV rms
Peak Transient Inputs
+3 V, −5 V Supply ±2.8 ±2.8 V p-p ±2.5 V Supply ±2.0 ±2.0 V p-p ±5 V Supply ±5.0 ±5.0 V p-p
Maximum Continuous
Nondestructive
Input Level (All Supply Voltages) ±12 ±12 V p-p Input Resistance 5.33 6.67 8 5.33 6.67 8 kΩ Input Offset Voltage ±0.5 ±0.2 mV
FREQUENCY RESPONSE
Bandwidth for 1% Additional
Error (0.09 dB)
VIN = 10 mV 14 14 kHz
VIN = 100 mV 90 90 kHz
VIN = 200 mV 130 130 kHz ±3 dB Bandwidth
VIN = 10 mV 100 100 kHz
VIN = 100 mV 900 900 kHz
VIN = 200 mV 1.5 1.5 MHz
OUTPUT CHARACTERISTICS3
Offset Voltage, VIN = COM
vs. Temperature ±10 ±10 V/°C
vs. Supply ±0.1 ±0.1 mV/V
Voltage Swing
+3 V, −5 V Supply 0.3 0 to 1.0 0.3 0 to 1.0 V
±5 V to ±16.5 V Supply 0.3 0 to 1.0 0.3 0 to 1.0 V
Output Impedance 8 10 12 8 10 12 kΩ
2, 3
3, 5
Min Typ Max Min Typ Max
OUT
1
AD636J AD636K
Unit
2
)
VavgV ×=
IN
±0.5 ± 1.0
±0.5
2
()
OUT
VavgV ×=
IN
±0.2 ± 0.5
±0.2
mV ± % of
ading
re mV ± % of
ading/°C
re mV ± % of
ading/V
re
mV ± % of
ading
re
mV
Rev. D | Page 3 of 16
AD636
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AD636J AD636K
Model
Min Typ Max Min Typ Max
dB OUTPUT
Error, VIN = 7 mV to 300 mV rms ±0.3
±0.5
±0.1
±0.2
Scale Factor −3.0 −3.0 mV/dB
Scale Factor Temperature
0.33 0.33 % of reading/°C
Coefficient
−0.033 −0.033 dB/°C I
for 0 dB = 0.1 V rms
REF
I
Range 1 50 1 50 A
REF
I
TERMINAL
OUT
I
Scale Factor 100 100 A/V rms
OUT
I
Scale Factor Tolerance −20 ±10 +20 −20 ±10 +20 %
OUT
2 4 8 2 4 8
Output Resistance 8 10 12 8 10 12 kΩ Voltage Compliance
−V (+V
to
S
− 2 V)
S
−V (+V
to
S
− 2 V)
S
V
BUFFER AMPLIFIER
Input and Output Voltage Range
−V (+V
to
S
− 2 V)
S
Input Offset Voltage, RS = 10 kΩ ±0.8 Input Bias Current 100
±2 300
to
−V
S
− 2 V)
(+V
S
±0.5 100
V
±1 300
Input Resistance 108 108 Ω Output Current
(+5 mA,
−130 A
)
(+5 mA,
−130 A
) Short-Circuit Current 20 20 mA Small Signal Bandwidth 1 1 MHz Slew Rate6 5 5 V/s
POWER SUPPLY
Voltage, Rated Performance +3, −5 +3, −5 V
Dual Supply +2, −2.5 ±16.5 +2, −2.5 ±16.5 V Single Supply 5 24 5 24 V
Quiescent Current7 0.80
1.00
0.80
1.00
TEMPERATURE RANGE
Rated Performance 0 +70 0 +70 °C Storage −55 +150 −55 +150 °C
TRANSISTOR COUNT 62 62
1
All minimum and maximum specifications are guaranteed. Specifications shown in boldface are tested on all production units at final electrical test and are used to
calculate outgoing quality levels.
2
Accuracy specified for 0 mV to 200 mV rms, dc or 1 kHz sine wave input. Accuracy is degraded at higher rms signal levels.
3
Measured at Pin 8 of PDIP (I
4
Error vs. crest factor is specified as additional error for a 200 mV rms rectangular pulse trim, pulse width = 200 µs.
5
Input voltages are expressed in V rms.
6
With 10 kΩ pull-down resistor from Pin 6 (BUF OUT) to −VS.
7
With BUF IN tied to COMMON.
), with Pin 9 tied to common.
OUT
Unit
dB
A
mV nA
mA
Rev. D | Page 4 of 16
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ABSOLUTE MAXIMUM RATINGS

Table 2.
Parameter Ratings
Supply Voltage
Dual Supply ±16.5 V
Single Supply 24 V Internal Power Dissipation 500 mW Maximum Input Voltage ±12 V Storage Temperature Range −55°C to +150°C Operating Temperature Range 0°C to 70°C Lead Temperature Range (Soldering 60 sec) 300°C ESD Rating 1000 V
1
10-Lead TO: θJA = 150°C/W.
14-Lead PDIP: θJA = 95°C/W.
1
PEAK
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.

ESD CAUTION

Rev. D | Page 5 of 16
AD636
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PIN CONFIGURATIONS AND FUNCTION DESCRIPTIONS

V
NC
–V
C
BUF OUT
BUF IN
1
IN
2
3
S
AD636
TOP VIEW
4
AV
(Not to Scale)
dB
5
6
7
NC = NO CO NNECT
14
+V
S
NC
13
NC
12
11
NC
COM
10
R
9
L
8
I
OUT
0787-003
Figure 2. 14-Lead SBDIP Pin Configuration
COM
Figure 3. 10-Pin TO-100 Pin Configuration
I
R
OUT
BUF IN
10
L
1
2
+V
S
Table 4. Pin Function Descriptions—10-Pin TO-100 Table 3. Pin Function Descriptions—14-Lead SBDIP
Pin No. Mnemonic Description Pin No. Mnemonic Description
1 VIN Input Voltage. 1 RL Load Resistor. 2 NC No Connection. 2 COM Common. 3 −V 4 C 5 dB
6 BUF OUT 7 BUF IN Buffer Input. 8 I 9 RL Load Resistor. 10 COM Common. 11, 12, 13 NC No Connection.
Negative Supply Voltage. 3 +VS Positive Supply Voltage.
S
Averaging Capacitor. 4 VIN Input Voltage.
AV
Log (dB) Value of the RMS Output
lta ge.
Vo Buffer Output.
RMS Output Current.
OUT
5 −VS Negative Supply Voltage. 6 C
Averaging Capacitor.
AV
7 dB Log (dB) Value of the RMS Output Voltage. 8 BUF OUT 9 BUF IN 10 I
RMS Output Current.
OUT
Buffer Output. Buffer Input.
14 +VS Positive Supply Voltage.
9
AD636
3
4
V
IN
BUF OUT
8
7
5
dB
6
C
AV
–V
S
00787-004
Rev. D | Page 6 of 16
AD636
erm
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APPLYING THE AD636

The input and output signal ranges are a function of the supply voltages as detailed in the specifications. The AD636 can also be used in an unbuffered voltage output mode by disconnecting the input to the buffer. The output then appears unbuffered across the 10 k resistor. The buffer amplifier can then be used for other purposes. Further, the AD636 can be used in a current output mode by disconnecting the 10 k resistor from the ground. The output current is available at Pin 8 (Pin 10 on the H package) with a nominal scale of 100 A per volt rms input, positive out.

STANDARD CONNECTION

The AD636 is simple to connect for the majority of high accuracy rms measurements, requiring only an external capacitor to set the averaging time constant. The standard connection is shown in
Figure 4. In this configuration, the AD636 measures the rms of the ac and dc level present at the input but shows an error for low frequency inputs as a function of the filter capacitor, C
, as shown in Figure 8. Therefore, if a
AV
4 F capacitor is used, the additional average error at 10 Hz is
0.1%, and at 3 Hz it is 1%. The accuracy at higher frequencies is according to specification. If it is desired to reject the dc input, a capacitor is added in series with the input, as shown in th
e capacitor must be nonpolar. If the AD636 is driven with
Figure 6;
power supplies with a considerable amount of high frequency ripple, it is advisable to bypass both supplies to ground with 0.1 F ceramic discs as near the device as possible. C
is an optional
F
output ripple filter.
C
F
erms
+V
–V
–V
C
+–
C
BUF OUT
BUF IN
V
IN
1
2
NC
AD636
S
3
AV
4
dB
5
6
+ BUF
7
NC = NO CONNECT
(OPTIO NAL)
ABSOLUTE
VALUE
SQUARER
DIVIDER
CURRENT
MIRROR
10k
10k
+V
14
13
12
11
10
9
8
S
+V
NC
NC
NC
COM
R
L
I
OUT
(OPTIO NAL)
+V
C
F
COM
+V
S
3
erms
R
2
V
IN
Figure 4. Standard RMS Connection
L
1
AD636
4
+
C
AV
I
OUT
10
10k
CURRENT
MIRROR
SQUARER
DIVIDER
ABSOLUTE
VALUE
–V
S
–V
BUF IN
9
+
BUF
BUF OUT
8
10k
7
dB
6
C
5
AV

OPTIONAL TRIMS FOR HIGH ACCURACY

If it is desired to improve the accuracy of the AD636, the external trims shown in t
rim the offset. The scale factor is trimmed by using R1 as shown. The insertion of R2 allows R1 to either increase or decrease the scale factor by ±1.5%.
Figure 5 can be added. R4 is used to
V
OUT
The trimming procedure is as follows:
Grou
nd the input signal, V
, and adjust R4 to give 0 V
IN
output from Pin 6. Alternatively, R4 can be adjusted to give the correct output with the lowest expected value of V
C
onnect the desired full-scale input level to V or a calibrated ac signal (1 kHz is the optimum frequency); then trim R1 to give the correct output from Pin 6, that is, 200 mV dc input should give 200 mV dc output. Of course, a ±200 mV peak-to-peak sine wave should give a 141.4 mV dc output. The remaining errors, as given in the specifications, are due to the nonlinearity.
C
AV
ABSOLUTE
VALUE
SQUARER
DIVIDER
CURRENT
MIRROR
+
BUF
10k
+
+V
S
14
13
12
11
10
9
10k
8
NC
NC
NC
COM
R
L
I
OUT
R2
154
+V
R3
470k
Offset Trims
SCALE
FACTOR
s
ADJUST
200
±1.5%
R1
–V
BUF OUT
V
OUT
–V
C
dB
BUF IN
NC
V
IN
1
2
S
3
AV
4
5
6
7
AD636
NC = NO CO NNECT
Figure 5. Optional External Gain and Output

SINGLE-SUPPLY CONNECTION

The applications in Figure 4 and Figure 5 assume the use of dual power supplies. The AD636 can also be used with only a single positive supply down to 5 V, as shown in Figure 6. Figure 6 is o
ptimized for use with a 9 V battery. The major limitation of this connection is that only ac signals can be measured because the input stage must be biased off ground for proper operation. This biasing is done at Pin 10; therefore, it is critical that no extraneous signals be coupled into this point. Biasing can be accomplished by using a resistive divider between +V
00787-005
ground. The values of the resistors can be increased in the interest of lowered power consumption, because only 1 µA of current flows into Pin 10 (Pin 2 on the H package).
Alternately, the COM pin of some CMOS ADCs provides a suitable a
rtificial ground for the AD636. AC input coupling requires only Capacitor C2 as shown; a dc return is not necessary because it is provided internally. C2 is selected for the proper low frequency break point with the input resistance of 6.7 k; for a cut-off at 10 Hz, C2 should be 3.3 F. The signal ranges in this connection are slightly more restricted than in the dual supply connection. The load resistor, R
, is necessary to provide current sinking capability.
L
, either dc
IN
+V
S
R4 500k
–V
S
OFFSET ADJUST
and
S
.
IN
00787-006
Rev. D | Page 7 of 16
AD636
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C2
3.3µF
V
IN
NONPOLARIZ ED
V
OUT
1kTO 10k
R
L
V
NC
–V
C
dB
BUF OUT
BUF IN
IN
1
2
S
3
AV
4
5
6
7
C
AV
+
ABSOLUTE
VALUE
AD636
SQUARER
DIVIDE R
CURRENT
MIRROR
+
BUF
NC = NO CONNECT
10k
10k
+V
S
14
0.1µF
NC
13
12
NC
NC
11
COM
10
R
9
I
OUT
8
20k
L
0.1µF
39k
Figure 6. Single-Supply Connection

CHOOSING THE AVERAGING TIME CONSTANT

The AD636 computes the rms of both ac and dc signals. If the input is a slowly varying dc voltage, the output of the AD636 tracks the input exactly. At higher frequencies, the average output of the AD636 approaches the rms value of the input signal. The actual output of the AD636 differs from the ideal output by a dc (or average) error and some amount of ripple, as demonstrated in
E
O
Figure 7.
IDEAL
E
O
DC ERROR = EO–EO(IDEAL)
100
0. 0
1%
E
10
(µF)
AV
1
VALUES FOR CAV AND
REQUIRED C
1% SETTLING TIME FOR
0.1 STATED % OF READING
AVERAGING ERRO R* ACCURACY ±20% DUE TO COMPONENT T OLERANCE *% dc ERROR + % RIPPLE (PEAK)
0.01 1 10 100 1k 10k 100k
0787-007
Figure 8. Error/Settling Time Graph for Use with the Standard RMS
1
0%
INPUT FREQ UENCY (Hz)
E
RROR
1% ERROR
Conne
0.1
ction
%
RROR
ERROR
100
10
1
0.1
0.01
MULTIPL Y READING BY 0.115
FOR 1% SETTLING TIME IN SECONDS
00787-009
The primary disadvantage in using a large CAV to remove ripple is that the settling time for a step change in input level is increased proportionately. bet of C
ween C
AV
and 1% settling time is 115 ms for each microfarad
AV
. The settling time is twice as great for decreasing signals
Figure 8 shows the relationship
as for increasing signals (the values in Figure 8 are for decreasing sig
nals). Settling time also increases for low signal levels, as
shown in
Figure 9.
10.0
DOUBLE-FREQ UENCY RIPPLE
AVERAGE E
O=EO
TIME
00787-008
Figure 7. Typical Output Waveform for Sinusoidal Input
The dc error is dependent on the input signal frequency and the value of C value of C
. Figure 8 can be used to determine the minimum
AV
which yields a given % dc error above a given
AV,
frequency using the standard rms connection.
The ac component of the output signal is the ripple. There are two ways to reduce the ripple. The first method involves using a large value of C
, a tenfold increase in this capacitance effects a tenfold
to C
AV
. Because the ripple is inversely proportional
AV
reduction in ripple. When measuring waveforms with high crest factors (such as low duty cycle pulse trains), the averaging time constant should be at least ten times the signal period. For example, a 100 Hz pulse rate requires a 100 ms time constant, which corresponds to a 4 F capacitor (time constant = 25 ms per F).
7.5
5.0
SETTLING TIME @ 200mV rms
SETTLING TIME RELATIVE TO
2.5
1.0
0
10mV 100mV
rms INPUT LEVEL
1V1mV
00787-010
Figure 9. Settling Time vs. Input Level
A better method for reducing output ripple is the use of a post-
lter. Figure 10 shows a suggested circuit. If a single-pole filter
fi
ed (C3 removed, R
is us 5 times the value of C
shorted), and C2 is approximately
X
, the ripple is reduced, as shown in
AV
Figure 11, and the settling time is increased. For example, with C
= 1 µF and C2 = 4.7 F, the ripple for a 60 Hz input is
AV
reduced from 10% of reading to approximately 0.3% of reading. The settling time, however, is increased by approximately a factor of 3. The values of C
and C2 can therefore be reduced
AV
to permit faster settling times while still providing substantial ripple reduction.
Rev. D | Page 8 of 16
AD636
V
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The 2-pole post filter uses an active filter stage to provide even greater ripple reduction without substantially increasing the settling times over a circuit with a 1-pole filter. The values of C
, C2, and C3 can then be reduced to allow extremely fast
AV
settling times for a constant amount of ripple. Caution should be exercised in choosing the value of C
, because the dc error
AV
is dependent upon this value and is independent of the post filter. For a more detailed explanation of these topics, refer to the
RMS-to-DC Conversion Application Guide, 2nd Edition.
IN
V
1
IN
NC
–V
S
–V
C
AV
+
+V
S
C
dB
BUF OUT
BUF IN
+
C2
ABSOLUTE
VALUE
2
AD636
3
SQUARER
DIVIDER
4
CURRENT
5
MIRROR
6
+
BUF
7
10k
10k
NC = NO CONNECT
Rx
Figure 10. 2-Pole Post Filter
10
p-p RIPPLE (ONE POLE) C
=1µF
AV
C2 = 4.7µF
10k
+V
S
14
+V
13
NC
12
NC
NC
11
COM
10
R
L
9
I
OUT
8
(FOR SINGLE POLE, SHORT Rx, REMOVE C3)
C3
+
OUT
V
rms
p-p RIPPLE C
=1µF
AV
(STANDARD CONNECT ION)
00787-011
A2. I1 drives one input of the squarer/divider, which has the transfer function:
2
I1
=
I4
I3
The output current, I4, of the squarer/divider drives the current
ror through a low-pass filter formed by R1 and the externally
mir connected capacitor, C
. If the R1, CAV time constant is much
AV
greater than the longest period of the input signal, then I4 is effectively averaged. The current mirror returns a current I3, which equals Avg. [I4], back to the squarer/divider to complete the implicit rms computation. Therefore,
2
I2
AvgI4 =
×=
I4
The current mirror also produces the output current, I which equals 2I4. I
can be used directly or converted to a
OUT
rmsI1
,
OUT
voltage with R2 and buffered by A4 to provide a low impedance voltage output. The transfer function of the AD636 thus results
V
= 2 R2 I rms = VIN rms
OUT
The dB output is derived from the emitter of Q3, because the voltage at this point is proportional to –log V
. Emitter follower,
IN
Q5, buffers and level shifts this voltage, so that the dB output voltage is zero when the externally supplied emitter current (I
) to Q5 approximates I3.
REF
CURRENTMIRROR
+V
14
S
1
DC ERROR OR RIPPLE (% of Reading)
0.1 10 100 1k 10k
DC ERROR C
=1µF
AV
(ALL FILTERS)
p-p RIPPLE (TWO POLE) C
= 1µF, C2 = C3 = 4.7µF
AV
FREQUENCY ( Hz)
00787-012
Figure 11. Performance Features of Various Filter Types

RMS MEASUREMENTS

AD636 Principle of Operation

The AD636 embodies an implicit solution of the rms equation that overcomes the dynamic range as well as other limitations inherent in a straightforward computation of rms. The actual computation performed by the AD636 follows the equation:
2
V
IN
×=
AvgrmsV
⎢ ⎢
Figure 12 is a simplified schematic of the AD636; it is sub
divided into four major sections: absolute value circuit (active rectifier), squarer/divider, current mirror, and buffer amplifier. The input voltage, V converted to a unipolar current I1, by the active rectifier A1,
rmsV
, which can be ac or dc, is
IN
V
IN

THE AD636 BUFFER AMPLIFIER

The buffer amplifier included in the AD636 offers the user additional application flexibility. It is important to understand some of the characteristics of this amplifier to obtain optimum performance. Figure 13 shows a simplified schematic of the buffer.
Because the output of an rms-to-dc converter is always positive, i
t is not necessary to use a traditional complementary Class AB output stage. In the AD636 buffer, a Class A emitter follower is used instead. In addition to excellent positive output voltage
Rev. D | Page 9 of 16
ABSOLUTE VAL UE/
VOLTAGE–CURRENT
CONVERTER
1
R3
10k
R4
20k
A1
8k
+
8k
I1
Q1
|
|V
IN
R4
A2
ONE-QU ADRANT
Figure 12. Simplifi
25k
I3
10µA
FS
A3
Q3
Q4Q2
SQUARER/
DIVIDER
ed Schematic
COM
10
R1
20µA FS
984
R
L
AVIOUT
BUF
IN
7
10k
C
AV
I
REF
BUFFER
A4
10k
R2
+V
S
dB
5
OUT
BUF
6
OUT
–V
3
S
00787-013
C
I4
Q5
AD636
V
V
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swing, this configuration allows the output to swing fully down to ground in single-supply applications without the problems associated with most IC operational amplifiers.
+
S
CURRENT
MIRROR
BUFFER
INPUT
5µA5µA
–V
S
Figure 13. AD636 Buffer Amplifier Simplified Schematic
When this amplifier is used in dual-supply applications as an input buffer amplifier driving a load resistance referred to ground, steps must be taken to ensure an adequate negative voltage swing. For negative outputs, current flows from the load resistor through the 40 k emitter resistor, setting up a voltage divider between −V
and ground. This reduced effective −VS,
S
limits the available negative output swing of the buffer. The addition of an external resistor in parallel with R voltage divider such that increased negative swing is possible.
Figure 14 shows the value of R
to −VS for several values of R
V
PEAK
R
increases the quiescent current of the buffer amplifier
EXTERNAL
by an amount equal to R current with no R
EXTERNAL
EXTERNAL
/−VS. Nominal buffer quiescent
EXT
is 30 µA at −VS = −5 V.
BUFFER OUTPUT
10k
R
E
40k
R
EXTERNAL
(OPTIONAL, SEE TEXT)
R
alters this
E
for a particular ratio of
. The addition of
LOAD
LOAD
00787-014
error. For example, note that a 1 V rms signal produces less than 1% of reading additional error up to 220 kHz. A 10 mV signal can be measured with 1% of reading additional error (100 µV) up to 14 kHz.
1V rms INPUT
1
200mV rms INPUT
200m
100mV rms INPUT
100m
30mV rms INPUT
30m
(V)
OUT
10m
V
10mV rms
INPUT
1m
1mV rms INPUT
0.1m 1k 10k 100k 1M 10M
1%
FREQUENCY (Hz)
10% ±3dB
00787-016
Figure 15. AD636 Frequency Response

AC MEASUREMENT ACCURACY AND CREST FACTOR (CF)

Crest factor is often overlooked in determining the accuracy of an ac measurement. Crest factor is defined as the ratio of the peak signal amplitude to the rms value of the signal (CF = V rms). Most common waveforms, such as sine and triangle waves, have relatively low crest factors (<2). Waveforms that resemble low duty cycle pulse trains, such as those occurring in switching power supplies and SCR circuits, have high crest factors. For example, a rectangular pulse train with a 1% duty cycle has a crest factor of 10 (CF = 1/√
η
).
/V
P
1.0
SUPPLY
/
PEAK
0.5
RATIO OF V
R
0
0 1k 10k 100k 1M
R
EXTERNAL
Figure 14. Ratio of Peak Negative Swing to −V
=6.7k
L
()
RL= 50k
R
=16.7k
L
vs. R
S
00787-015
EXTERNAL
for Several Load Resistances

FREQUENCY RESPONSE

The AD636 uses a logarithmic circuit to perform the implicit rms computation. As with any log circuit, bandwidth is proportional to signal level. The solid lines in Figure 15 r
epresent the frequency response of the AD636 at input levels from 1 mV to 1 V rms. The dashed lines indicate the upper frequency limits for 1%, 10%, and ±3 dB of reading additional
Rev. D | Page 10 of 16
Figure 16 is a curve of reading error for the AD636 for a 200 mV r
ms input signal with crest factors from 1 to 7. A rectangular pulse train (pulse width 200 s) was used for this test because it is the worst-case waveform for rms measurement (all the energy is contained in the peaks). The duty cycle and peak amplitude were varied to produce crest factors from 1 to 7 while maintaining a constant 200 mV rms input amplitude.
0.5
T
V
P
0
E
0
200µs
O
ŋ
= DUTY CYCL E =
CF = 1/
ŋ
(rms) = 200mV
E
IN
–0.5
INCREASE IN ERRO R (% of Readi ng)
–1.0
1234567
CREST FACTOR
Figure 16. Error vs. Crest Factor
200µs
T
00787-017
AD636
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A COMPLETE AC DIGITAL VOLTMETER

Figure 17 shows a design for a complete low power ac digital voltmeter circuit based on the AD636. The 10 M input attenuator allows full-scale ranges of 200 mV, 2 V, 20 V, and 200 V rms. Signals are capacitively coupled to the AD636 buffer amplifier, which is connected in an ac bootstrapped configuration to minimize loading. The buffer then drives the 6.7 k input impedance of the AD636. The COM terminal of the ADC provides the false ground required by the AD636 for single­supply operation. An AD589 1.2 V reference diode is used to provide a stable 100 mV reference for the ADC in the linear rms mode; in the dB mode, a 1N4148 diode is inserted in series to provide correction for the temperature coefficient of the dB scale factor. Calibration of the meter is done by first adjusting offset trimmer R17 for a proper zero reading, and then adjusting the R13 for an accurate readout at full scale.
Calibration of the dB range is accomplished by adjusting R9 fo
r the desired 0 dB reference point, and then adjusting R14 for the desired dB scale factor (a scale of 10 counts per dB is convenient).
1.2 V AD589 band gap reference, and finally back to the negative side of the battery via R10. This sets ground at 1.2 V + 3.18 V (250 A × 12.7 k) = 4.4 V below the positive battery terminal and
5.0 V (250 A × 20 k) above the negative battery terminal. Bypass capacitors, C3 and C5, keep both sides of the battery at a low ac impedance to ground. The AD589 band gap reference establishes the 1.2 V regulated reference voltage, which together with R3 and trimming Potentiometer R4, sets the 0 dB reference current, I
REF
.

Performance Data

0 dB Reference Range = 0 dBm (770 mV) to −20 dBm (77 mV) rms
0 dBm = 1 mW in 600  Input Range (at I Input Impedance = approximately 10 V
Operating Range = +5 V dc to +20 V dc
SUPPLY
I Accuracy with 1 kHz sine wave and 9 V dc supply:
= 1. 8 mA typical
QUIESCENT
0 dB to −40 dBm ± 0.1 dBm
m to −50 dBm ± 0.15 dBm
0 dB +10 dBm to −50 dBm ± 0.5 dBm
= 770 mV) = 50 dBm
REF
10
Total power supply current for this circuit is typically 2.8 mA usin
g a 7106-type ADC.

A LOW POWER, HIGH INPUT, IMPEDANCE dB METER

The portable dB meter circuit combines the functions of the AD636 rms converter, the AD589 voltage reference, and a A776 low power operational amplifier (see Figure 18). This m
eter offers excellent bandwidth and superior high and low level accuracy while consuming minimal power from a standard 9 V transistor radio battery.
In this circuit, the built-in buffer amplifier of the AD636 is us
ed as a bootstrapped input stage increasing the normal 6.7 k
input Z to an input impedance of approximately 10

Circuit Description

The input voltage, VIN, is ac-coupled by C4 while R8, together with D1 and D2, provide high input voltage protection.
The buffer’s output, Pin 6, is ac-coupled to the rms converter’s
put (Pin 1) by capacitor C2. Resistor R9 is connected between
in the buffer’s output, a Class A output stage, and the negative output swing. Resistor R1 is the amplifier’s bootstrapping resistor.
With this circuit, single-supply operation is made possible by s
etting ground at a point between the positive and negative sides of the battery. This is accomplished by sending 250 A from the positive battery terminal through R2, then through the
10
.

Frequency Response ±3 dBm

Input
0 dBm = 5 Hz to 380 kHz
−10 dB
m = 5 Hz to 370 kHz
−20 dBm = 5 Hz to 240 kHz
−30 dBm = 5 Hz to 100 kHz
−40 dBm = 5 Hz to 45 kHz
−50 dBm = 5 Hz to 17 kHz

Calibration

First, calibrate the 0 dB reference level by applying a 1 kHz sine wave from an audio oscillator at the desired 0 dB amplitude. This can be anywhere from 0 dBm (770 mV rms − 2.2 V p-p) to −20 dBm (77 mV rms − 220 mV p-p). Adjust the I calibration trimmer for a zero indication on the analog meter.
Then, calibrate the meter scale factor or gain. Apply an input sig
nal −40 dB below the set 0 dB reference and adjust the scale
factor calibration trimmer for a 40 A reading on the analog meter.
The temperature compensation resistors for this circuit can be p
urchased from Micro-Ohm Corporation, 1088 Hamilton Rd.,
Duarte, CA 91010, Part #Type 401F, 2 k ,1% + 3500 ppm/°C.
REF
Rev. D | Page 11 of 16
AD636
V
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47k
IN
9M
900k
90k
10k
200mV
R1
R2
R3
R4
COM
2V
20V
200V
C3
0.02µF
SIGNAL
INPUT
10%
1N6263
R5
1W
1N6263
C4
0.1µF
47k
R8
1W
D2
1N4148
R6
1M
6.8µF
20k
1N4148
D1
D1
+
C4
2.2µF
V
IN
1
ABSOLUTE
NC
2
–V
S
3
C
AV
dB
BUF IN
4
5
6
7
+
BUF OUT
R7
D4
VALUE
AD636
SQUARER
DIVIDER
CURRENT
MIRROR
+
BUF
NC = NO CO NNECT
10k
10k
Figure 17. Portable, High-Z Input, RMS
+
C1
3.3µF
R1
1M
6.8µF
10k
C2
+
R9
V
IN
1
ABSOLUT E
2
NC
–V
S
3
C
AV
4
dB
5
BUF OUT
ALL RESISTORS 1/4W 1% METAL F ILM UNLE SS OTHERW ISE STAT ED EXCEPT *WHICH IS 2k +3500ppm 1% TC RESI STOR.
BUF IN
6
7
VALUE
AD636
SQUARER
DIVIDE R
CURRENT
MIRROR
+
BUF
10k
NC = NO CONNECT
10k
+V
S
14
13
NC
12
NC
11
NC
COM
10
R
L
9
I
OUT
8
+
C7
6.8µF
R9 100k 0dB SET
R10 20k
D3
1.2V
AD589
LIN
dB
LIN
dB
R8
2.49k
R11
10k
LIN
SCALE
R15
1M
D2
1N4148
R12 1k
R13 500
DPM and dB Meter Circuit
12.7k
C3 10µF
250µA
C5 10µF
R10
20k
+4.4V
R2
+1.2V
R3
5k
+
AD589J
*R7
2k
R6
100
C6
0.1µF
+4.7V
+V
S
14
13
NC
+
12
NC
NC
11
COM
10
R
L
9
+
I
8
OUT
LIN
dB
R14 10k dB SCALE
C6
0.01µF
R4 500k I
REF
ADJUST
100µA
2
3
+V
S
+V
CONVERTER
REF HI
REF LO
COM
HI
ANALOG
LO
–V
S
–V
ON/OFF
SCALE FACTOR
+ –
7
µA776
8
+
4
R11 820k 5%
DD
3-1/2 DIG IT 7106 TYPE
A/D
IN
SS
+ –
9V
ADJUST
R5
10k
0–50µA
6
+V
1µF
–V
DD
+
SS
BATTERY
DISPLAY
LXD 7543
00787-019
ON
9V
3-1/2
DIGIT
LCD
OFF
+
00787-018
Figure 18. Low Power, High Input Impedance dB Meter
Rev. D | Page 12 of 16
AD636
C
R
.
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OUTLINE DIMENSIONS

0.005 (0.13) MIN
PIN 1
0.200 (5.08) MAX
0.200 (5.08)
0.125 (3.18)
0.023 (0.58)
0.014 (0.36)
ONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FO REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN
Figure 19. 14-Lead Side-Brazed Ceram
Dimensions shown in inches and (millimeters)
REFERENCE PLANE
0.080 (2.03) MAX
14
1
0.100 (2.54) BSC
0.765 (19.43) MAX
0.070 (1.78)
0.030 (0.76)
8
0.310 (7.87)
0.220 (5.59)
7
0.320 (8.13)
0.060 (1.52)
0.015 (0.38)
0.150 (3.81) MIN
SEATING PLANE
0.290 (7.37)
ic Dual In-Line Package [SBDIP]
(D-14)
0.015 (0.38)
0.008 (0.20)
0.185 (4.70)
0.165 (4.19)
0.370 (9.40)
0.335 (8.51)
0.335 (8.51)
0.040 (1.02) MAX
0.050 (1.27) MAX
0.500 (12.70) MIN
0.021 (0.53)
0.016 (0.40)
0.305 (7.75)
BASE & SEATING PLANE
DIMENSIONS PER JEDEC STANDARDS MO-006-AF CONTROLL ING DIMENS IONS ARE IN INCHES; MILLIMETER DI MENSIONS (IN PARENTHESES) ARE ROUNDED-OF F INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRI ATE FOR USE IN DES IGN.
0.115 (2.92)
BSC
0.230 (5.84) BSC
0.160 (4.06)
0.110 (2.79)
5
4
3
2
6
7
8
0.045 (1.14)
9
0.025 (0.65)
1
10
36° BSC
0.034 (0.86)
0.025 (0.64)
022306-A
Figure 20. 10-Pin Metal Header Package [TO-100]
(H-10)
Dim
ensions shown in inches and (millimeters)

ORDERING GUIDE

Model Temperature Range Package Description Package Option
AD636JD 0°C to +70°C 14-Lead SBDIP D-14 AD636JDZ1 0°C to +70°C 14-Lead SBDIP D-14 AD636KD 0°C to +70°C 14-Lead SBDIP AD636KDZ 0°C to +70°C 14-Lead SBDIP
1
AD636JH 0°C to +70°C 10-Pin TO-100 H-10 AD636JHZ
1
0°C to +70°C 10-Pin TO-100 H-10 AD636KH 0°C to +70°C 10-Pin TO-100 AD636KHZ
1
Z = Pb-free part.
1
0°C to +70°C 10-Pin TO-100 H-10
D-14 D-14
H-10
Rev. D | Page 13 of 16
AD636
www.BDTIC.com/ADI
NOTES
Rev. D | Page 14 of 16
AD636
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NOTES
Rev. D | Page 15 of 16
AD636
www.BDTIC.com/ADI
NOTES
©2006 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. C00787-0-11/06(D)
Rev. D | Page 16 of 16
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